Broadcast signal transmitter/receiver and broadcast signal transmitting/receiving method

ABSTRACT

A broadcast signal transmitter according to one embodiment of the present invention comprises: an input signal generating unit which generates a first input signal and a second input signal; a MIMO encoder which performs MIMO processing of the first input signal and the second input signal to output a first transmission signal and a second transmission signal; and a first OFDM generator and a second OFDM generator which performs OFDM modulation of the first transmission signal and OFDM modulation of the second transmission signal. The MIMO processing applies a MIMO matrix to the first input signal and the second input signal. The MIMO matrix changes phases using a phase rotation matrix, and adjusts power of the first input signal and second input signal using parameter a, wherein the parameter is set to different values in accordance with the modulation types of the first input signal and second input signal.

FIELD OF THE INVENTION

The present invention relates to a method for transceiving broadcastsignals and an apparatus for transceiving broadcast signals, and moreparticularly, to a method for transceiving broadcast signals, which canenhance data transmission efficiency and is compatible with conventionalmethods for transceiving broadcast signals, and a transceiving apparatusthereof.

BACKGROUND ART

As analog broadcasting will soon end, a variety of technologies fortransmitting and receiving digital broadcast signals has been developed.Digital broadcast signals can transmit a greater capacity of video/audiodata than analog broadcast signals, and can include a variety ofoptional data in addition to video/audio data.

A digital broadcast system can provide High Definition (HD) images,multi-channel sound, and a variety of optional services. However, datatransmission efficiency for high capacity data transmission, robustnessof transmitting and receiving networks, and flexibility of networks inconsideration of mobile receiving equipment are problems that shouldstill be improved.

DETAILED DESCRIPTION OF THE INVENTION Technical Problem

A technical object of the present invention is to provide a method andapparatus for transceiving broadcast signals, which can receive digitalbroadcast signals without error even under an indoor environment orusing mobile receiving equipment.

Another technical object of the present invention is to provide a methodand apparatus for transceiving broadcast signals, which can maintaincompatibility with a conventional broadcast system in addition toachieving the above described objects.

Technical Solution

The object of the present invention can be achieved by providing amethod for transmitting a broadcast signal including: generating a firstinput signal and a second input signal; outputting a first transmissionsignal and a second transmission signal by MIMO-encoding the first inputsignal and the second input signal; and OFDM-modulating each of thefirst transmission signal and the second transmission signal, whereinthe MIMO encoding applies a MIMO matrix to the first input signal andthe second input signal, the MIMO matrix changes a phase using a phaserotation matrix, and adjusts power of the first input signal and thesecond input signal using a parameter (a), where the parameter (a) isset to a different value according to a modulation type of the firstinput signal and the second input signal.

Effects of the Invention

According to the present invention, in a digital broadcast system, it ispossible to enhance data transmission efficiency and increase robustnessin terms of transmission and reception of broadcast signals, by virtueof provision of a MIMO system.

Further, according to the present invention, in a digital broadcastsystem, it is possible to decode MIMO receiving signals efficientlyusing MIMO processing of the present invention even under a diversebroadcast environment.

In addition, according to the present invention, a broadcast systemusing MIMO of the present invention can achieve the above describedadvantages while maintaining compatibility with a conventional broadcastsystem not using MIMO.

Further, according to the present invention, it is possible to provide amethod and apparatus for transceiving broadcast signals, which canreceive digital broadcast signals without error even under an indoorenvironment or using mobile reception equipment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a broadcast signal transmitter using MIMO according to anembodiment of the present invention.

FIG. 2 shows a broadcast signal receiver according to an embodiment ofthe present invention.

FIG. 3 shows an additional frame structure based on PLP according to anembodiment of the present invention.

FIG. 4 shows an additional frame structure based on FEF according to anembodiment of the present invention.

FIGS. 5 (A) and 5 (B) show a process of generating a P1 symbol in orderto perceive an additional frame according to an embodiment of thepresent invention.

FIG. 6 shows a conceptual diagram of a broadcast signal transmittingmethod according to an embodiment of the present invention.

FIG. 7 shows a conceptual diagram of a broadcast signal transmittingmethod according to another embodiment of the present invention.

FIG. 8 shows a broadcast signal transmitted by a national broadcastsystem with a MIMO system applied using SVC.

FIG. 9 shows a structure of a P1 symbol and AP1 symbol according to anembodiment of the present invention.

FIG. 10 shows a P1 symbol detection module according to an embodiment ofthe present invention.

FIG. 11 shows an AP1 symbol detection module according to an embodimentof the present invention.

FIG. 12 shows an input processor of a broadcast signal transmitteraccording to an embodiment of the present invention.

FIG. 13 shows a mode adaption module implementing a plurality of PLP asan input processor according to an embodiment of the present invention.

FIG. 14 shows a stream adaption module implementing a plurality of PLPas an input processor according to an embodiment of the presentinvention.

FIG. 15 shows a BICM encoder according to an embodiment of the presentinvention.

FIG. 16 shows a BICM encoder according to another embodiment of thepresent invention.

FIG. 17 shows a frame builder encoder according to an embodiment of thepresent invention.

FIG. 18 shows an OFDM generator according to an embodiment of thepresent invention.

FIG. 19 shows an OFDM demodulator according to an embodiment of thepresent invention.

FIG. 20 shows a frame demapper according to an embodiment of the presentinvention.

FIG. 21 shows a BICM decoder according to an embodiment of the presentinvention.

FIG. 22 shows a BICM decoder according to another embodiment of thepresent invention.

FIG. 23 shows an output processor according to an embodiment of thepresent invention.

FIG. 24 shows an output processor according to another embodiment of thepresent invention.

FIG. 25 illustrates a MIMO transceiving system according to anembodiment of the present invention.

FIG. 26 is a BER/SNR chart showing a performance difference between a GCscheme and an SM scheme using an outer code according to an embodimentof the present invention.

FIG. 27 is a BER/SNR chart showing a performance difference between a GCscheme and an SM scheme according to modulation schemes and a code rateof an outer code according to an embodiment of the present invention.

FIG. 28 illustrates a data transceiving method based on MIMOtransmission of SM scheme in a channel environment according to anembodiment of the present invention.

FIG. 29 illustrates input signals and a transmitted/received signal onwhich a MIMO encoding method according to an embodiment of the presentinvention has been performed.

FIG. 30 is a BER/SNR chart showing the performance of a MIMO encodingmethod according to a first embodiment of the present invention.

FIG. 31 is a capacity/SNR chart showing the performance of the MIMOencoding method according to the first embodiment of the presentinvention in an uncorrelated channel.

FIG. 32 is a capacity/SNR chart showing the performance of the MIMOencoding method according to the first embodiment of the presentinvention in a fully correlated channel.

FIG. 33 shows constellations when a sub-set of GS is used as a MIMOencoding matrix and in the case of the first embodiment of the presentinvention.

FIG. 34 is a capacity/SNR chart showing performances when the sub-set ofGS is used as the MIMO encoding matrix and in the case of the firstembodiment of the present invention.

FIG. 35 illustrate the relationship between a Euclidean distance andhamming distance in the constellations when a sub-set of GS is used as aMIMO encoding matrix and in the case of the first embodiment of thepresent invention.

FIG. 36 illustrates input signals and a transmitted/received signal onwhich a MIMO encoding method according to a second embodiment of thepresent invention has been performed.

FIG. 37 illustrates a MIMO encoding method according to a thirdembodiment of the present invention.

FIG. 38 illustrates input signals and a transmitted/received signal onwhich the MIMO encoding method according to the third embodiment of thepresent invention has been performed.

FIG. 39 is a capacity/SNR chart showing performances of the MIMOencoding methods according to the present invention.

FIG. 40 is another capacity/SNR chart showing performances of the MIMOencoding methods according to the present invention.

FIG. 41 is a capacity/SNR chart showing performances according tocombinations of modulation schemes in the MIMO encoding method accordingto the third embodiment of the present invention.

FIG. 42 is a capacity/SNR chart showing performances according tochannel correlation when QPSK+QPSK MIMO transmission scheme is used inthe MIMO encoding method according to the third embodiment of thepresent invention.

FIG. 43 is a capacity/SNR chart showing performances according tochannel correlation when QPSK+16-QAM MIMO transmission scheme is used inthe MIMO encoding method according to the third embodiment of thepresent invention.

FIG. 44 is a capacity/SNR chart showing performances according tochannel correlation when 16-QAM+16-QAM MIMO transmission scheme is usedin the MIMO encoding method according to the third embodiment of thepresent invention.

FIG. 45 illustrates input signals and transmission signals on which aMIMO encoding method according to a fourth embodiment of the presentinvention has been performed.

FIG. 46 illustrates input signals and transmission signals on which aMIMO encoding method according to a sixth embodiment of the presentinvention has been performed.

FIG. 47 is a flowchart illustrating a method of transmitting a broadcastsignal according to an embodiment of the present invention.

FIG. 48 is a flowchart illustrating a method of receiving a broadcastsignal according to an embodiment of the present invention.

BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, although the preferred embodiments of the present inventionwill be described in detail with reference to the accompanying drawingsand contents as described with relation to the accompanying drawings, itis to be understood that the present invention is not limited to theembodiments.

Various technologies have been introduced to increase transmissionefficiency and to perform robust communication in a digital broadcastsystem. One of such technologies is a method of using a plurality ofantennas at a transmitting side or a receiving side. This method may beclassified into a Single-Input Single-Output (SISO) scheme in whichtransmission is performed through a single antenna and reception isperformed through a single antenna, a Single-Input Multi-Output (SIMO)scheme in which transmission is performed through a single antenna andreception is performed through multiple antennas, a Multi-InputSingle-Output (MISO) scheme in which transmission is performed throughmultiple antennas and reception is performed through a single antenna,and a Multi-Input Multi-Output (MIMO) scheme in which transmission isperformed through multiple antennas and reception is performed throughmultiple antennas. Although the multiple antennas may be exemplified by2 antennas for ease of explanation in the following description, thedescription of the present invention may be applied to systems that use2 or more antennas.

The SISO scheme corresponds to a general broadcast system that uses 1transmission antenna and 1 reception antenna. The SIMO schemecorresponds to a broadcast system that uses 1 transmission antenna and aplurality of reception antennas.

The MISO scheme corresponds to a broadcast system that uses a pluralityof transmission antennas and 1 reception antenna to provide transmitdiversity. An example of the MISO scheme is an Alamouti scheme. In theMISO scheme, it is possible to receive data through 1 antenna withoutperformance loss. Although a reception system can receive the same datathrough a plurality of reception antennas in order to improveperformance, this case will be described as belonging to MISO cases inthis specification.

A MIMO scheme corresponds to a broadcast system that uses a plurality oftransmit (Tx) antennas and a plurality of receive (Rx) antennas toprovide transmission/reception (Tx/Rx) diversity and high transmissionefficiency. In the MIMO scheme, signals are processed in different waysin time and space dimensions and a plurality of data streams istransmitted through parallel paths that simultaneously operate in thesame frequency band to achieve diversity effects and high transmissionefficiency.

The performance of a system that employs the MIMO technology depends oncharacteristics of a transmission channel. The efficiency of such asystem is high, especially, when the system has independent channelenvironments. That is, the performance of the system that employs theMIMO technology may improve when channels of all antennas ranging fromantennas of the transmitting side and antennas of the receiving side areindependent channels that have no correlation to each other. However, ina channel environment in which the correlations between channels oftransmission and reception antennas are very high as in a line-of-sight(LOS) environment, the performance of the system that employs the MIMOtechnology may be significantly reduced or the system may not be able tooperate.

In addition, if the MIMO scheme is applied to a broadcast system thatuses the SISO and MISO schemes, it is possible to increase datatransmission efficiency. However, in addition to the above problems,there is a need to maintain compatibility to allow a receiver having asingle antenna to receive services. Accordingly, the present inventionsuggests a method for solving such existing problems.

In addition, the present invention can provide a broadcast signaltransmitter/receiver and a broadcast transmission and reception methodfor a conventional terrestrial broadcast system and a system that cantransmit and receive additional broadcast signals (or enhanced broadcastsignals), for example, mobile broadcast signals, while sharing an RFfrequency band with a terrestrial broadcast system such as DVB-T2.

To accomplish this, in the present invention, it is possible to use avideo coding method having scalability in which a basic video componentwhich has low image quality although it is robust to a communicationenvironment and an extended video component which is slightly weak to acommunication environment although it can provide a high-quality imagecan be distinguishably transmitted. Although the present invention willbe described with reference to SVC as a video coding method havingscalability, the present invention may be applied to any other videocoding methods. Embodiment of the present invention will be described inmore detail with reference to the drawings.

A broadcast signal transmitter and receiver of the present invention canperform MISO processing and MIMO processing on a plurality of signalsthat are transmitted and received through a plurality of antennas. Thefollowing is a description of a broadcast signal transmitter andreceiver that performs signal processing on 2 signals that aretransmitted and received through 2 antennas.

FIG. 1 shows a broadcast signal transmitter using MIMO according to anembodiment of the present invention.

As shown in FIG. 1, the broadcast signal transmitter according to thepresent invention may include an input processor 101100, an inputprocessing module 101200, a Bit Interleaved Coded Modulation (BICM)encoder 101300, a frame builder 101400, and an OrthogonalFrequency-Division Multiplexing (OFDM) generator (or transmitter)101500. The broadcast signal transmitter according to the presentinvention may receive a plurality of MPEG-TS streams or a General StreamEncapsulation (GSE) stream (or GS stream).

The input processor 101100 may generate a plurality of PLPs (physicallayer pipes) on a service basis in order to give robustness to aplurality of input streams, i.e., a plurality of MPEG-TS streams or GSEstreams.

PLPs are data units that are identified in the physical layer.Specifically, a PLP is data having the same physical layer attributewhich is processed in the transmission path and may be mapped on a cellby cell basis in a frame. In addition, a PLP may be considered aphysical layer Time Division Multiplexing (TDM) channel that carries oneor a plurality of services. Specifically, a path through which such aservice is transmitted is transmitted or a stream identifiable in thephysical layer which is transmitted through the path is referred to as aPLP.

Thereafter, the input processing module 101200 may generate a Base Band(BB) frame including a plurality of generated PLPs. The BICM module101300 may add redundancy to the BB frame to correct an error in atransmission channel and may interleave PLP data included in the BBframe.

The frame builder 101400 may accomplish a transmission frame structureby mapping the plurality of PLPs to a transmission frame and addingsignaling information thereto. The OFDM generator 101500 may demodulateinput data from the frame builder according to OFDM to divide the inputdata into a plurality of paths such that the input data is transmittedthrough a plurality of antennas.

FIG. 2 shows a broadcast signal receiver according to an embodiment ofthe present invention.

As shown in FIG. 2, the broadcast signal receiver may include an OFDMdemodulator 107100, a frame parser 107200, a BICM decoder 107300, and anoutput processor 107400. The OFDM demodulator 107100 may convert signalsreceived through a plurality of receive antennas into signals in thefrequency domain. The frame parser 107200 may output PLPs for anecessary service from among the converted signals. The BICM decoder107300 may correct an error generated according to a transmissionchannel. The output processor 107400 may perform procedures necessary togenerate output TSs or GSs. Here, dual polarity signals may be input asinput antenna signals and one or more streams may be output as the TXsor GSs.

FIG. 3 shows an additional frame structure based on PLP according to anembodiment of the present invention.

As shown in FIG. 3, a frame according to an embodiment of the presentinvention may include a preamble area and a data area. The preamble areamay include a P1 symbol and a P2 symbol and the data area may include aplurality of data symbols. The P1 symbol may transmit P1 signalinginformation and P2 symbol may transmit L1-signaling information.

In this case, a preamble symbol may be additionally allocated to thepreamble. This additional preamble symbol is referred to as anAdditional Preamble 1 (AP1). In an embodiment of the present invention,one or more AP1 symbols may be added to a frame in order to improvedetection performance of a mobile broadcast signal under very low SNR ortime-selective fading conditions. AP1 signaling information transmittedthrough the AP1 symbol may include an additional transmission parameter.

AP1 signaling information according to an embodiment of the presentinvention includes pilot pattern information in a frame. Thus, accordingto an embodiment of the present invention the broadcast signal receiverdoes not transmit P2 symbol, if L1 signaling information is spread indata symbols of the data area, pilot pattern information can bediscovered by using the AP1 signaling information before L1 signalinginformation in the data area is decoded.

Also, if the L1-signaling information in the data area of a frame isspread, AP1 signaling information can include information necessary forthe broadcast signal receiver to decode signaling information spread ina frame of the data area. According to the present invention, a preamblearea of a frame includes a P1 symbol, more than one AP1 symbols, andmore than one P2 symbols. And the data area comprises a plurality ofdata symbols, also known as data OFDM symbol. A P2 symbol is optionaland whether it is inserted is determined by signaling AP1 signalinginformation through AP1 symbols according to an embodiment of thepresent invention.

In an embodiment of the present invention, a P1 insertion module in theOFDM generator OFDM generator 101500 of the broadcast signal transmittermay insert the P1 symbol and the AP1 symbol into every symbol. That is,the P1 insertion module may insert 2 or more preamble symbols into everyframe. In another embodiment, an AP1 insertion module may be addeddownstream of (or next to) the P1 insertion module and the AP1 insertionmodule may insert the AP1 symbol. If 2 or more preamble symbols are usedas in the present invention, there are advantages in that robustness toburst fading that may occur in a mobile fading environment is furtherincreased and signal detection performance is also improved.

The P1 symbol may transmit P1 signaling information associated with abasic transmission parameter and transmission type and a correspondingpreamble identifier and the receiver may detect the frame using the P1symbol. A plurality of P2 symbols may be provided and may carry L1signaling information and signaling information such as a command PLP.The L1 signaling information may include L1-pre signaling informationand L1-post signaling information, the L1-pre signaling information mayinclude information necessary to receive and decode the L-post signalinginformation. Also, the L-post signaling information may includeparameters necessary for the receiver to encode PLP data. As shown inFIG. 3, the L-post signaling information may be located next to L1-presignaling information.

The L-post signaling information may include a configurable block, adynamic block, an extension block, a cyclic redundancy check (CRC)block, and an L padding block.

The configurable block may include information equally applied to onetransmission frame and the dynamic block may include characteristicinformation corresponding to a currently transmitted frame.

The extension block may be used when the L1-post signaling informationis extended, and the CRC block may include information used for errorcorrection of the L1-post signaling information and may have 32 bits.The padding block may be used to adjust sizes of informationrespectively included in a plurality of encoding blocks to be equal whenthe L1-post signaling information is transmitted while being dividedinto the encoding blocks and has a variable size.

The common PLP may include network information such as a NIT(NetworkInformation Table) or PLP information and service information such as anSDT(Service Description Table) or an EIT(Event Information Table). Thepreamble of the present invention may include only the P1 symbol, theL1-pre signaling information, and the L1-post signaling information ormay include all of the P1 symbol, the L1-pre signaling information, theL1-post signaling information, and the common PLP according to designerintention. A plurality of data symbols located next to the P1 symbol mayinclude a plurality of PLPs. The plurality of PLPs may include audio,video, and data TS streams and PSI/SI information such as a ProgramAssociation Table (PAT) and a Program Map Table (PMT). In the presentinvention, a PLP that transmits PSI/SI information may be referred to asa base PLP or a signaling PLP. The PLPs may include a type-1 PLP that istransmitted through one sub-slice per frame and a type-2 PLP that istransmitted through two sub-slices per frame. The plurality of PLPs maytransmit one service and may also transmit service components includedin one service. When the PLPs transmit service components, thetransmitting side may transmit signaling information which indicatesthat the PLPs transmit service components.

In addition, additional data (or an enhanced broadcast signal) inaddition to basic data may be transmitted through a specific PLP whilesharing an RF frequency band with the conventional terrestrial broadcastsystem according to an embodiment of the present invention. In thiscase, the transmitting side may define a system or a signal that iscurrently transmitted through signaling information of the P1 symboldescribed above. The following description is given with reference tothe case in which the additional data is video data. That is, as shownin FIG. 3, PLP M1 112100 and PLP (M1+M2) 112200 which are type 2 PLPsmay be transmitted while including additional video data. In addition,in the present invention, a frame that transmits such additional videodata may be referred to as an additional frame and a frame thattransmits basic data may be referred to as a basic frame (or T2 frame).

In addition, a frame that can transmit not only additional data but alsodata associated with a new broadcast system different from theconventional terrestrial broadcast system may be referred to as anadditional frame. In this case, a frame that transmits a conventionalterrestrial broadcast may be referred to as a terrestrial broadcastframe and an additional frame may transmit additional data or basic dataassociated with the new broadcast system.

FIG. 4 illustrates a structure of an additional frame based on FEFaccording to an embodiment of the present invention.

Specifically, FIG. 4 shows the case in which a Future Extension Frame(FEF) is used in order to transmit additional video data. In the presentinvention, a frame that transmits basic video data may be referred to asa basic frame and an FEF that transmits additional video data may bereferred to as an additional frame.

FIG. 4 shows structures of superframes 11100 and 113200 in each of whicha basic frame and an additional frame are multiplexed. Frames 113100-1to 113100-n that are not shaded from among frames included in thesuperframe 113100 are basic frames and shaded frames 113120-1 and113120-2 are additional frames.

FIG. 4(A) shows the case in which the ratio of basic frames toadditional frames is N:1. In this case, the time required for thereceiver to receive a next additional frame 113120-2 after receiving oneadditional frame 113120-1 may correspond to N basic frames.

FIG. 4(B) shows the case in which the ratio of basic frames toadditional frames is 1:1. In this case, the proportion of additionalframes in the superframe 113200 may be maximized and therefore theadditional frames may have a structure very similar to that of the basicframes in order to maximize the extent of sharing with the basic frames.In addition, in this case, the time required for the receiver to receivea next additional frame 113210-2 after receiving one additional frame113210-1 corresponds to 1 basic frame 113220 and therefore thesuperframe period is shorter than that of FIG. 4(A).

FIGS. 5(A) and 5(B) illustrate a P1 symbol generation procedure foridentifying additional frames according to an embodiment of the presentinvention.

In the case in which additional video data is transmitted throughadditional frames which are distinguished from basic frames as shown inFIG. 4, there is a need to transmit additional signaling information forenabling the receiver to identify and process an additional frame. Anadditional frame of the present invention may include a P1 symbol fortransmitting such additional signaling information and the P1 symbol maybe referred to as a new_system_P1 symbol. This new_system_P1 symbol maybe different from a P1 symbol that is used in a conventional frame and aplurality of new_system_P1 symbols may be provided. In an embodiment,the new_system_P1 symbol may be located before a first P2 symbol in apreamble area of the frame.

In the present invention, a P1 symbol of a conventional frame may bemodified and used to generate the minimum Hamming distance. The presentinvention suggests a method in which a minimum Hamming distance isgenerated by modifying the structure of the P1 symbol of theconventional frame or is generated by changing the symbol generator114100 that generates symbols.

FIG. 5(A) shows the structure of the P1 symbol of the conventionalframe. In the present invention, the structure of the P1 symbol of theconventional frame shown in FIG. 5(A) may be modified to generate aminimum Hamming distance. In this case, the minimum Hamming distance maybe generated by changing a frequency displacement f_SH for the prefixand postfix of the conventional P1 symbol or changing the length(specifically, the size of T_P1C or T_P1B) of the P1 symbol. However, inthe case in which the minimum Hamming distance is generated by modifyingthe structure of the P1 symbol, there is a need to appropriately modifyparameters (the sizes of T_P1C and T_PIB and f_SH) used in the P1 symbolstructure.

FIG. 5(B) shows the P1 symbol generator that generates P1 symbols. Inthe present invention, the P1 symbol generator shown in FIG. 5(B) may bemodified to generate a minimum Hamming distance. In this case, a minimumHamming distance may be generated using a method which changes thedistribution of active carriers used for a P1 symbol in a CDS tablemodule 114110, an MSS module 114120, and a C-A-B structure module 114130included in the P1 symbol generator (for example, a method in which theCDS table module 114110 uses a different Complementary Set of Sequence(CSS)) or a method which changes a pattern for information that istransmitted through a P1 symbol (for example, a method in which the MSSmodule 114120 uses a different Complementary Set of Sequence (CSS)).

In addition, the AP1 symbol of the present invention described abovewith reference to FIG. 3 may be generated through the proceduredescribed above with reference to FIG. 5.

In addition, the present invention proposes a MIMO system using scalablevideo coding (SVC). SVC is a video coding method developed to cope witha variety of terminals and communication environments and variations inthe terminals and communication environments. SVC can code a videohierarchically such that desired definition is generated and transmitadditional video data having a base layer from which video data about animage having basic definition can be restored and an enhancement layerfrom which an image having higher definition can be restored.Accordingly, a receiver can acquire the basic definition image byreceiving and decoding only the video data of the base layer, or obtainthe higher definition image by decoding the video data of the base layerand the video data of the enhancement layer according to characteristicsthereof. In the following description, the base layer can include videodata corresponding to the base layer and the enhancement layer caninclude video data corresponding to the enhancement layer. In thefollowing, video data may not be a target of SVC, the base layer caninclude data capable of providing a fundamental service including basicvideo/audio/data corresponding to the base layer, and the enhancementlayer can include data capable of providing a higher service includinghigher video/audio/data corresponding to the enhancement layer.

The present invention proposes a method of transmitting the base layerof SVC through a path through which signals can be received according toSISO or MISO using SVC and transmitting the enhancement layer of SVCthrough a path through which signals can be received according to MIMOin the broadcast system of the present invention. That is, the presentinvention provides a method by which a receiver having a single antennaacquires an image with basic definition by receiving the base layerusing SISO or MISO and a receiver having a plurality of antennasacquires an image with higher definition by receiving the base layer andthe enhancement layer using MIMO.

A description will be given of a method of transmitting the MIMObroadcast data including the base layer and the enhancement layer inassociation with terrestrial broadcast frames for transmittingterrestrial broadcast signals.

(1) Method of Transmitting MIMO Broadcast Data Using Predetermined PLP

It is possible to transmit the MIMO broadcast data included in apredetermined PLP while distinguishing the predetermined PLP from a PLPincluding terrestrial broadcast data. In this case, the predeterminedPLP is used to transmit the MIMO broadcast data, and signalinginformation for describing the predetermined PLP may be additionallytransmitted to prevent an error in the conventional receiving system. Inthe following, the predetermined PLP including the MIMO broadcast datamay be referred to as a MIMO broadcast PLP and the PLP including theterrestrial broadcast data may be referred to as a terrestrial broadcastPLP.

As MIMO broadcast data may not be implemented in a terrestrial broadcastreceiver, it is necessary to have additional information for signallingto distinguish terrestrial PLP and MIMO broadcast PLP. In this case,signaling can use a reserved field in the L1 signaling information ofthe terrestrial broadcast system. When a plurality of antennas are usedfor transmitting MIMO broadcast data on the transmitting side, theterrestrial broadcast data can be transmitted by MISO. The presentinvention, in order to perceive PLP, utilizes L1-post signalinginformation.

(2) Method of Transmitting MIMO Broadcast Data Using Predetermined Frame

It is possible to include the MIMO broadcast data generated as describedabove in a predetermined frame and to transmit the predetermined frameincluding the MIMO broadcast data while distinguishing the predeterminedframe from a terrestrial broadcast frame. In this case, thepredetermined frame is used to transmit the MIMO broadcast data, andsignaling information for describing the predetermined frame may beadditionally transmitted to prevent an error in the conventionalreceiving system.

(3) Method of Transmitting MIMO Broadcast PLP Using TerrestrialBroadcast Frame and MIMO Broadcast Frame

PLPs including MIMO broadcast data may be transmitted through aterrestrial broadcast frame and a MIMO broadcast frame. Since a MIMObroadcast PLP may be present in the terrestrial broadcast frame (orbasic frame), distinguished from the above-mentioned embodiments, it isnecessary to signal the relationship between connected PLPs present inthe terrestrial broadcast frame and the MIMO broadcast frame. To achievethis, the MIMO broadcast frame may also include L1 signalinginformation, and information about the MIMO broadcast PLP present in thebroadcast frame may be transmitted along with L1 signaling informationof the terrestrial broadcast frame.

MIMO broadcast PLP data in different frames are connected by using PLPfields including L-post signaling information. According to anembodiment of the present invention, the receiving system includes asL1-post signaling information PLP_ID information, PLP+TYPE information,PLP_PAYLOAD_TYPE information, PLP_GROYP_ID information, uses thoseinformation to check the PLP connection between MIMO broadcast PLP data.It then acquires services by continuously decoding desired MIMObroadcast PLP data.

The terrestrial broadcast PLP in the terrestrial broadcast frames can betransmitted as a preset mode and also as mentioned a new mode to supportthe MIMO system can be transmitted. According to an embodiment of thepresent invention, the MIMO broadcast PLP in the terrestrial broadcastframes as a base layer can be transmitted by MISO or SISO method andMIMO broadcast PLP in MIMO broadcast frames as an enhancement layer canbe transmitted by the MIMO method.

FIG. 6 shows a conceptual diagram for a method of transmitting broadcastsignals.

As shown in FIG. 6, terrestrial broadcast data and MIMO broadcast datain frame units can be distinctively transmitted. The FEF length of aMIMO broadcast frame (FEF) can be allocated in between terrestrialbroadcast frames in an FEF interval. In this case, MIMO system data canco-exist in a frequency band within the terrestrial broadcast system,and malfunction can be prevented by the broadcast signal receiverperceiving a frame through L1 signaling and ignoring MIMO broadcastframes. In that case, the MIMO system can use some of the thruput by FEFrelated parameters such as FEF_TYPR, FEF_LENGTH, FEF_INTERVAL defined bythe L1-post signaling information.

FIG. 7 shows a conceptual diagram for a broadcast signal transmittingmethod according to another embodiment of the present invention.

FIG. 7 indicates, as shown in the method 3, transmitting the broadcastsignals of the MIMO broadcast system in terrestrial broadcast system.The MIMO broadcast services (MIMO broadcast service 1˜n) encodes eachSVC encoder (18010, 18020) through a base layer and enhancement layer.Scheduler&BICM (Bit Interleaved Coding and Modulation) module (18030)allocates the base layers of the MIMO broadcast services with theterrestrial broadcast frames and the enhancement layers with MIMOencoders (18040, 18050). The enhancement layers encodes by each MIMOencoder (18040, 18050) and transmits to the MIMO broadcast frame of theMIMO broadcast system. The base layers are transmitted in theterrestrial broadcast frames and in that case, SISO or MISO supported bythe terrestrial broadcast system.

When broadcast signals including the terrestrial broadcast frames andthe MIMO broadcast frames, as mentioned in the method 1 and 3, signalinginformation is created and the terrestrial broadcast receiver perceivesterrestrial broadcast PLP in the terrestrial broadcast frames. Thus, thereceiver can acquire the terrestrial broadcast services withoutmalfunctioning. Also, the MIMO broadcast receiver can acquire andprovide the MIMO broadcast service corresponding to the base layer onlyby the terrestrial broadcast frame. It can acquire and provide the MIMObroadcast service corresponding to the base layer and enhancement layerby acquiring the MIMO broadcast PLP of the terrestrial broadcast frameand MIMO broadcast frame of the MIMO broadcast frame.

The MIMO broadcast PLP in the terrestrial broadcast frame can only betransmitted by MISO/MIMO. In that case, the MIMO broadcast PLP, as thesystem demands, can include a code rate of new error correctioncodes(such as ¼, ⅓, ½), and new time interleaving mode and can onlytransmit to a base layer.

The MIMO broadcast PLP of the MIMO broadcast frame includes PLP of theSISO, MISO, and MIMO methods. In that case, PLP of the SISO/MISO methodsor a base layer in a carrier can be transmitted and PLP of the MIMOmethod or the carrier can transmit the enhancement layer. The rate ofPLP of the SISO/MISO methods, or carrier and PLP of the MIMO method, orcarrier can be varied from 0 to 100%. The ract can be determined foreach frame accordingly.

FIG. 8 shows broadcast signals transmitted by a broadcast system beingapplied by a MIMO system using a SVC.

FIG. 8 shows a broadcast signal that allocates terrestrial data and MIMObroadcast data to a frame or PLP by using the SVC and generating a baseand enhancement layer.

FIG. 8 A shows a broadcast signal transmitted by a broadcast systembeing applied by a MIMO transmitting system by using the SVC.

The broadcast system in FIG. 8 A transmits broadcast signals including aterrestrial broadcast frame and MIMO broadcast frame. The MIMO broadcastPLP in FIG. 8 A can exist in a terrestrial broadcast frame or a MIMObroadcast frame. The MIMO broadcast PLP in the terrestrial broadcastframe as a base layer can be transmitted by the SISO or MISO method andthe MIMO broad cast PLP in the MIMO broadcast frame as an enhancementlayer can be transmitted by the SISO, MISO, or MIMO method.

FIG. 8B shows a broadcast signal being applied by a MIMO transmittingsystem using a SVC.

In FIG. 8B, the broadcast system transmits broadcast signals includingthe terrestrial broadcast frame and the MIMO broadcast frame. The MIMObroadcast PLP in FIG. 8 B only exists in the MIMO broadcast frame. Inthat case, the MIMO broadcast PLP includes PLP with a base layer and PLPwith an enhancement layer. The PLP with the base layer can betransmitted by the SISO or MISO method, and the PLP with the enhancementlayer can be transmitted by the SISO, MISO, or MIMO method. The rate ofthe PLP with base layer and the PLP with enhancement layer can be variedfrom 0 to 100%.

FIG. 8 C shows a broadcast signal transmitted by a broadcast systembeing applied by a MIMO transmitting system using a SVC.

The broadcast system of FIG. 8 C transmits broadcast signals includingterrestrial broadcast frames and MIMO broadcast frames. The MIMObroadcast data exists only in the MIMO broadcast frame. But, as opposedto FIG. 8 B, a base layer and an enhancement layer are not transmittedby PLP but carriers.

Various technologies are introduced to improve transmission efficiencyand perform robust communication in a digital broadcast system. One ofthe technologies is a method of using a plurality of antennas at atransmitting side or a receiving side. This method may be divided intoSISO(Single-Input Single-Output), SIMO(Single-Input Multi-Output),MISO(Multi-Input Single-Output) and MIMO(Multi-Input Multi-Output).While multiple antennas are described as two antennas in the following,the present invention is applicable to systems using two or moreantennas.

In an embodiment, MIMO can use spatial multiplexing (SM) and Golden code(GC) schemes, which will be described in detail.

A modulation scheme in broadcast signal transmission may be representedas M-QAM (Quadrature Amplitude Modulation) in the following description.That is, BPSK (Binary Phase Shift Keying) can be represented by 2-QAMwhen M is 2 and QPSK (Quadrature Phase Shift Keying) can be representedby 4-QAM when M is 4. M can indicate the number of symbols used formodulation.

A description will be given of a case in which a MIMO system transmitstwo broadcast signals using two transmit antennas and receives twobroadcast signals using two receive antennas as an example.

FIG. 9 shows an exemplary structure of a P1 symbol and an exemplarystructure of an AP1 symbol according to an embodiment of the presentinvention.

P1 symbol is generated by having each of a front portion and an endportion of an effective (or valid) symbol copied, by having a frequencyshift performed as much as +f_(sh), and by having the frequency-shiftedcopies respectively positioned at a front portion (C) and an end portion(B) of the effective symbol (A). In the present invention, the C portionwill be referred to as a prefix, and the B portion will be referred toas a postfix. More specifically, P1 symbol is configured of a prefixportion, an effective symbol portion, and a postfix portion.

In the same manner, AP1 symbol is generated by having each of a frontportion and an end portion of an effective (or valid) symbol copied, byhaving a frequency shift performed as much as −f_(sh), and by having thefrequency-shifted copies respectively positioned at a front portion (F)and an end portion (E) of the effective symbol (D). In the presentinvention, the F portion will be referred to as a prefix, and the Eportion will be referred to as a postfix. More specifically, AP1 symbolis configured of a prefix portion, an effective symbol portion, and apostfix portion.

Herein, the two frequency-shift values +f_(sh), −f_(sh), which are usedin the P1 symbol and the AP1 symbol, may have the same absolute valueyet be given opposite signs. More specifically, the frequency-shift isperformed in opposite directions. And, the lengths C and F, which arecopied to the front portion of the effective symbol, may be set to havedifferent values. And, the lengths B and E, which are copied to the endportion of the effective symbol, may be set to have different values.Alternatively, the lengths C and F may be set to have different values,and the lengths B and E may be set to have the same value, or viceversa. According to another embodiment of the present invention, aneffective symbol length of the P1 symbol and an effective symbol lengthof the AP1 symbol may be differently determined. And, according to yetanother embodiment of the present invention, a CSS (Complementary SetSequence) may be used for tone selection and data scrambling within theAP1 may be scrambled by AP1.

According to the embodiment of the present invention, the lengths of Cand F, which are copied to the front portion of the effective (or valid)symbol, may be set to have different values, and the lengths of B and E,which are copied to the end portion of the effective (or valid) symbol,may also be set to have different values.

The C,B,F,E lengths according to the present invention may be obtainedby using Equation 1 shown below.

Length of C(T _(C))={Length of A(T _(A))/2+30}

Length of B(T _(B))={Length of A(T _(A))/2−30}

Length of E(T _(F))={Length of D(T _(D))/2+15}

Length of E(T _(E))={Length of D(T _(D))/2−15}  [Expression 1]

As shown in Equation 1, P1 symbol and AP1 symbol have the same frequencyshift value. However, each of the P1 symbol and the AP1 symbol are givenopposite signs. Additionally, in order to determine the lengths of C andB, the present invention determines an offset value being added to orsubtracted from a value corresponding to the length of A (T_(A))/2. And,in order to determine the lengths of F and E, the present inventiondetermines an offset value being added to or subtracted from a valuecorresponding to the length of D (T_(D))/2. Herein, each of the offsetvalues is set up differently. According to the embodiment of the presentinvention, the offset value of P1 symbol is set to 30, and the offsetvalue of AP1 symbol is set to 15. However, the values given in theabove-described examples are merely exemplary. And, therefore, it willbe apparent that the corresponding values may easily be varied orchanged by anyone skilled in the art. Thus, the present invention willnot be limited only to the values presented herein.

According to the present invention, by generating AP1 symbol and an AP1symbol to configure the structure shown in FIG. 9, and by inserting thegenerated symbols to each signal frame, the P1 symbol does not degradethe detection performance of the AP1 symbol, and, conversely, the AP1symbol does not degrade the detection performance of the P1 symbol.Additionally, the detection performance of the P1 symbol is almostidentical to the detection performance of the AP1 symbol. Furthermore,by configuring the symbols so that the P1 symbol and the AP1 symbol havesimilar symbol structures, the complexity level of the receiver may bereduced.

At this point, the P1 symbol and the AP1 symbol may be transmittedconsecutively, or each of the symbols may be allocated to differentpositions within the signal frame and may then be transmitted. And, incase the P1 symbol and AP1 symbol are each allocated to a differentposition within the signal frame, so as to be transmitted, a high timediversity effect may be gained with respect to the preamble symbol.According to the embodiment of the present invention, the P1 symbol andthe AP1 symbol are consecutively transmitted. In that case, the AP1symbol, according to FIG. 3, transmits information necessary fordecoding signaling information spread in a pilot pattern or a frame of adata area. It can be generated in FIG. 5.

FIG. 10 shows an exemplary structure of a P1 symbol detector accordingto an embodiment of the present invention.

The P1 symbol detector may be included in the OFDM demodulator(107100)explained in FIG. 2.

Herein, the P1 symbol detector may also be referred to as a C-A-Bpreamble detector. The P1 symbol detector may include down shifter(307101), 1^(st) conjugator (307103) and 2^(nd) delayer (307106).

The down shifter (307101) performs inverse modulation by multiplyinge^(−j2πf) ^(SH′) by the input signal. When inverse modulation isperformed by the down shifter (307101), the signal beingfrequency-shifted and inputted is recovered to the original signal. Theinverse modulated signal may be outputted to a 1^(st) delayer (307102)and a 2^(nd) conjugator (307107).

The 1^(st) delayer (307102) delays the inverse-modulated signal by alength of part C (T_(C)) and then outputs the delayed signal to the1^(st) conjugator (307103). The 1^(st) conjugator (307103) performscomplex-conjugation on the signal, which is delayed by a length of partC (T_(C)). Then, the 1^(st) conjugator (307103) multiplies the inputsignal by the complex-conjugated signal, thereby outputting theprocessed signal to a 1^(st) filter (307104). The 1^(st) filter (307104)uses a running average filter having the length of T_(R)=T_(A), so as toremove (or eliminate) any excessively and unnecessarily remainingmodulation elements, thereby outputting the processed signal to a 3^(rd)delayer (307105). The 3^(rd) delayer (307105) delays the filtered signalby a length of part A (i.e., effective (or valid) symbol) (T_(A)), so asto output the delayed signal to a multiplier (307109).

The 2^(nd) delayer (307106) delays the input signal by a length of partB (T_(B)) and then outputs the delayed signal to the 2^(nd) conjugator(307107). The 2^(nd) conjugator (307107) performs complex-conjugation onthe signal, which is delayed by a length of part B (T_(B)). Then, the2^(nd) conjugator (307107) multiplies the complex-conjugated signal byan inverse-modulated signal, thereby outputting the processed signal toa 2^(nd) filter (307108). The 2^(nd) filter (307108) uses a runningaverage filter having the length of T_(R)=T_(A), so as to remove (oreliminate) any excessively and unnecessarily remaining modulationelements, thereby outputting the processed signal to the multiplier(307109).

The multiplier (307109) multiplies the output of the 2^(nd) filter(307109) by a signal, which is delayed by a length of part A (T_(A)).Thus, a P1 symbol may be detected from each signal frame of the receivedbroadcast signal.

Herein, the length of part C (T_(C)) and the length of part B (T_(B))may be obtained by applying Equation 1 shown above.

FIG. 11 shows an exemplary structure of an AP1 symbol detector accordingto an embodiment of the present invention.

The AP1 symbol detector may be included in the OFDM demodulator(107100)explained in FIG. 2.

Herein, the AP1 symbol detector may also be referred to as an F-D-Epreamble detector. The AP1 symbol detector may include down shifter(308101), 1^(st) conjugator (308103) and 2^(d) delayer (308106). The AP1symbol detector may receive a signal inputted to broadcast signalreceiver or a signal outputted from the P1 symbol detector explained inFIG. 10.

The up-shifter (308101) performs inverse modulation by multiplyinge^(j2πf) ^(SH′) by the input signal. When inverse modulation isperformed by the up-shifter (308101), the signal being frequency-shiftedand inputted is recovered to the original signal. More specifically, theup-shifter (308101) of FIG. 47 has the same structure as thedown-shifter (307101) of the P1 symbol detector (306601). However, thefrequency direction of each inverse modulation process is completelyopposite to one another. The signal that is inverse modulated by theup-shifter (308101) may be outputted to a 1^(st) delayer (308102) and a2nd conjugator (308107).

The 1^(st) delayer (308102) delays the inverse-modulated signal by alength of part F (T_(F)) and then outputs the delayed signal to the1^(st) conjugator (308103). The 1^(st) conjugator (308103) performscomplex-conjugation on the signal, which is delayed by a length of partF (T_(F)). Then, the 1^(st) conjugator (308103) multiplies the inputsignal by the complex-conjugated signal, thereby outputting theprocessed signal to a 1^(st) filter (308104). The 1^(st) filter (308104)uses a running average filter having the length of T_(R)=T_(D), so as toremove (or eliminate) any excessively and unnecessarily remainingmodulation elements, thereby outputting the processed signal to a 3^(rd)delayer (308105). The 3^(rd) delayer (308105) delays the filtered signalby a length of part D (i.e., effective (or valid) symbol) (T_(D)), so asto output the delayed signal to a multiplier (308109).

The 2^(nd) delayer (308106) delays the input signal by a length of partE (T_(E)) and then outputs the delayed signal to the 2^(nd) conjugator(308107). The 2^(nd) conjugator (308107) performs complex-conjugation onthe signal, which is delayed by a length of part E (T_(E)). Then, the2^(nd) conjugator (308107) multiplies the complex-conjugated signal byan inverse-modulated signal, thereby outputting the processed signal toa 2^(nd) filter (308108). The 2^(nd) filter (308108) uses a runningaverage filter having the length of T_(R)=T_(D), so as to remove (oreliminate) any excessively and unnecessarily remaining modulationelements, thereby outputting the processed signal to the multiplier(308109).

The multiplier (308109) multiplies the output of the 2^(nd) filter(308109) by a signal, which is delayed by a length of part D (T_(D)).Thus, an AP1 symbol may be detected from each signal frame of thereceived broadcast signal. Herein, the length of part F (T_(F)) and thelength of part E (T_(E)) may be obtained by applying Equation 1 shownabove.

As shown in FIG. 3, a frame according to an embodiment of the presentinvention comprises a preamble area and a data area. The preamble arecomprises a P1 and P2 and there can be a plurality of data symbols inthe data area. Also, as the designer intends, there can be an AP1 in thepreamble area.

Then, P1 signaling information is transmitted by the P1 symbol, the AP1signaling information is transmitted by the AP1 symbol, and L1-pre andL1-post signaling information is transmitted by the P2 symbol.

An embodiment of a broadcast signal transmitter or receiver for MIMOprocessing is as follows.

The broadcast signal transmitter comprises as shown in FIG. 1 an inputprocessor 101200, a BICM encoder 101300, a frame builder 101400, and anOFDM generator 101500. Also, the broadcast signal receiver, as shown inFIG. 2, comprises an OFDM demodulator 107100, a frame demapper 107200, aBICM decoder 107300, and an output processor 1073400.

The input processor 101200 of the broadcast signal transmitter executesFEC encoding for transmitting data in a form of block. The BICM encoder101300 performs encoding for correcting errors. The frame builder 101400performs mapping data in a frame, and the OFDM generator 101500 performsOFDM demodulating in the frame-mapped data into symbol units andtransmit the data. Devices in the broadcast signal receiver can performreverse-functioning corresponding to the counterpart devices in thetransmitter.

The present invention suggests a broadcast signal transmitter orreceiver that independently applies MISO or MIMO processing for each PLPfrom a plurality of PLP inputs. According to the present invention, thepresent invention can effectively adjust the quality of service (QOS) orservices from PLP in a physical layer.

Four embodiments for performing MISO/MISO processing in a plurality ofsignals from the transmitter and receiver through a plurality ofantennas are as follows. Individual embodiments can be distinguishedfrom each other according to whether MISO/MIMO processing for each PLPis processed or according to the position of MISO/MIMO processing. Abrief description of individual embodiments is as follows.

A first embodiment is about a broadcast signal transmitter or acorresponding receiver independently performing MISO or MIMO processingfor each PLP data input during a BICM encoding process.

A second embodiment is about another broadcast signal transmitter or acorresponding receiver independently performing MISO or MIMO processingfor each PLP data input during a BICM encoding process.

A third embodiment is about a broadcast signal transmitter or acorresponding receiver independently performing MISO or MIMO processingfor mapped PLP data input during a OFDM generating process.

A fourth embodiment is about a broadcast signal transmitter or acorresponding receiver independently performing MISO or MIMO processingfor each PLP data input during a BICM encoding process, wherein an OFDMgenerator performs MISO processing in MISO PLP data and L1-signalinginformation.

In more detail, the BICM encoder of the broadcast signal transmitteraccording to the first embodiment performs MISO encoding or MIMOencoding in PLP data after constellation-mapping, cell interleaving, andtime interleaving. Also, the BICM decoder of the broadcast signaltransmitter according to the first embodiment can reverse the wholeprocess. According to the second embodiment, the BICM encoder of thebroadcast signal transmitter according to the second embodiment performsMISO encoding or MIMO encoding in PLP data after constellation-mapping,and then performs cell interleaving and time interleaving. Also, theBICM decoder of the broadcast signal transmitter according to the secondembodiment can reverse the whole process.

According to the third embodiment, the OFDM generator of the broadcastsignal transmitter performs MISO or MIMO encoding in PLP datatransmitted from a frame builder. In addition, an OFDM demodulator ofthe broadcast signal receiver according to a third embodiment of thepresent invention may perform a reverse process of the OFDM generator ofthe broadcast transmitter.

According to the fourth embodiment, the BICM encoder of the broadcastsignal transmitter according to the fourth embodiment performs MISOencoding or MIMO encoding in PLP data after time interleaving orconstellation-mapping. Also, the OFDM generator of the broadcast signaltransmitter performs MISO encoding in MISO PLP data for MISO processingand L-signaling information. The BICM decoder of the broadcast signalreceiver and the OFDM demodulator of the broadcast signal transmitteraccording to the fourth embodiment can reverse the whole process.

A broadcast signal transmitter/receiver according to each embodiment isas follows. The broadcast signal transmitter/receiver can perform MIMOprocessing for a plurality of signals through a plurality of antennas.The broadcast signal transmitter/receiver with two signals by twoantennas is described below.

FIG. 12 and FIG. 13 show an input process that the broadcast signaltransmitter comprises in common. Further description is as follows.

FIG. 12 shows an input processor of the broadcast signal transmitteraccording to an embodiment.

The input process 101200 in FIG. 1 is shown as an embodiment in FIG. 13performing only one PLP. The input processor in FIG. 12 comprises a modeadaptation module 601100 and a stream adaptation module 601200. The modeadaptation module 601100 further comprises an input interface module601110, a CRC-8 encoder 601120 and a BB header insertion module 601130,wherein a stream adaptation module 1020 comprises a padding insertionmodule 601210 and a BB scrambler 601220.

The input interface module 601110 in the input processor performing asingle PLP performs mapping by distinguishing the input bit stream in alogical unit to perform FEC (BCH/LDPC) encoding at the end of the BICMencoder. The CRC-9 encoder 601120 performs CRC encoding in the mappedbit stream and a BB header insertion module 1050 inserts a BB header inthe data field. In that case, the BB header includes all adaptation type(TS/GS/IP) information, user packet length information, and data fieldlength.

Also, if the input data does not have a BB frame for FEC encoding, thestream adaptation block 601200 generates a padding insertion unit and aPseudo Random Binary Sequence (PRBS) and includes a BB scrambler 601220randomizing data computed by the PRBS and XOR. Such a move by the BBscrambler 601220 can ultimately lower the Peak-to-Average Power Ratio ofthe OFDM-modulated signal.

FIG. 13 shows a mode adaptation module implementing a plurality of PLPas an input processor according to an embodiment of the presentinvention.

FIG. 13 shows a mode adaptation module as an input processor of thebroadcast signal transmitter performing a plurality of PLP. The modeadaptation module in FIG. 14 comprises a plurality of input interfacemodules 602100 performing mode adaptation for each PLP in parallel, aninput stream synchronizer 602200, a compensating delay module 602300,null packet deletion module 602400, a CRC-0 encoder 602500, and a BBheader insertion unit 602600. The description of the input interfacemodule 6021000, the CRC-8 encoder 602500 and the BB header insertionunit 602600 is omitted.

The input stream synchronizer 602200 inserts timing informationnecessary for restoring input stream clock reference information (ISCR),transport stream (TS) or generic stream (GS). The compensating delaymodule 602300 synchronizes a group of PLP based on the timinginformation. The null packet deletion module (602400) deletes nullpacket that is unnecessarily transmitted and inserts the number of thedeleted null packets based on the deleted position.

FIG. 14 shows a stream adaptation module implementing a plurality of PLPas an input processor according to an embodiment of the presentinvention.

The stream adaptation module in FIG. 14 receives PLP-based data in whichmode adaptation of FIG. 13 was performed from the mode adaptation moduleof FIG. 13, such that it can perform stream adaptation as shown in thefollowing description.

The scheduler 603100 performs scheduling for the MIMO transmittingsystem using a plurality of antennas including dual polarity andgenerates parameters for a demultiplexer, a cell interleaver, a timeinterleaver. Also, the scheduler 603100 transmits L1-dynamic signalinginformation for the current frame besides in-band signaling, andperforms cell mapping based on the scheduling.

A plurality of a 1-frame delay module 603200 executing a plurality ofPLP delays one frame so that scheduling information of the next framefor in-band signaling can be included in the current frame. A pluralityof in-band signaling/padding insertion module inserts L-dynamicsignaling information to the delayed data. Also, if there is any roomfor padding, the in-band signaling/padding insertion module 603300inserts padding bits and in-band signaling information into the paddingarea. And, the BB scrambler 603400 generates a pseudo random binarysequence (PRBS) as shown in FIG. 29 and randomizes the data by computingthe PRBS with XOR.

The stream adaption module in FIG. 14 generates L-signaling informationtransmitted by the preamble symbol of the frame or the spread datasymbol. Such L1-signaling information includes L1-pre signalinginformation and L1-post signaling information. The L1-pre signalinginformation includes parameters necessary for performing the L1-postsignaling information and static L1-signaling information, and theL1-post signaling information includes the static L1-signalinginformation and dynamic L1-signaling information. The L-signalinggenerator 603500 can transmit the generated L1-pre signaling informationand L-post signaling information. The transmitted L1-pre signalinginformation and L1-post signaling information is scrambled by each BBscramble 603600, 603700. Also, according to another embodiment, the L1signaling generator 603500 transmits L1-signaling information havingL1-pre signaling and L1-post signaling information and scramblesL1-signaling information transmitted by one BB scrambler.

FIG. 15 to FIG. 18 shows a structure block of a broadcast signaltransmitter according to an embodiment. Further description is asfollows.

FIG. 15 shows a BICM encoder according to an embodiment of the presentinvention.

The BICM encoder shown in FIG. 15 is an embodiment of the BICM encoder101300 in FIG. 1.

The BICM encoder according to the first embodiment performsbit-interleaving in a plurality of PLP data after performinginput-processing, L1-pre signaling information, and L1-post signalinginformation, and encoding for correcting errors.

Also, the BICM encoder independently performs MISO or MIMO encoding inPLP data. In addition, the BICM encoder according to a first embodimentof the present invention may perform MISO encoding and MIMO encodingupon completion of constellation mapping.

The BICM encoder in FIG. 15 includes a first BICM encoding block 607100performing MISO encoding in PLP data, a second BICM encoding block607200 performing MIMO encoding in PLP data, and a third BICM encodingblock 607300 performing MIMO encoding in signaling information. Thethird BICM encoding block 604300 performing MIMO encoding in signalinginformation. However, as the signaling information includes informationnecessary for restoring PLP data in a frame from the receiver, morerobustness is required between the transmitter and receiver compared toPLP data. Thus, an embodiment of the present invention is the MISOprocess performing the signaling information. The description of dataperforming process for each block is as follows.

First, the first BICM encoding block 604100 includes a BICM encoder604100, a FEC (Forward Error Correction) encoder 604110, abit-interleaver 604120, a first demultiplexer 604130, a constellationmapper 604140, a MISO encoder 604150, a cell interleaver 604160-1,604160-2 and a time interleaver 604170-1, 604170-2.

The FEC encoder 604110 performs BCH encoding and LDPC encoding in PLPdata after performing input processing with redundancy to correctchannel errors from the receiver. The bit-interleaver 604120 prepares tohave robustness for bust errors by performing bit-interleaving in theFEC-encoded PLP data by each FEC block unit. In that case, the bitinterleaver can perform bit interleaving by using two FEC block units.When using two FEC blocks, a pair of cell units may be generated fromtwo different FEC blocks in the frame-builder. Thus, the broadcastsignal receiver may improve the reception by ensuring the diversity ofFEC blocks.

A first demultiplexer 604130 can perform demultiplexing in thebit-interleaved PLP data into one FEC block unit. In that case, thefirst demultiplexer 604130 uses two FEC blocks and performsdemultiplexing. When using the two blocks, pairs of cells in the framebuilder may be generated from different FEC blocks. Thus, the receivercan improve reception by ensuring the diversity of FEC blocks.

The constellation mapper 604140 performs mapping in thebit-demultiplexed PLP data into symbol units. In that case, theconstellation mapper 604140 can rotate a certain angle depending on themodulation type. The rotated constellation mappers can be expressed inI-phase (In-phase) and Q-phase (Quadrature-phase), and the constellationmappers can delay only the Q-phase for a certain value. Then, theconstellation mapper 604140 performs re-mapping in the In-phase elementwith the delayed Q-phase element.

The MISO encoder 604150 performs MISO encoding by using MISO encodingmatrix in the time-interleaved PLP data and transmits MISO PLP datathrough two routes (STx_k, STx_k+1). The present invention includes anOSTBC (Orthogonal Space-Time Block Code)/OSFBC(Orthogonal SpaceFrequency Block Code/Alamouti code) as an embodiment of a MISO encodingmethod.

The cell interleaver 604160-1, 604160-2 performs interleaving in there-mapped data into cell units, and the time interleaver 604170-1,604170-2 performs interleaving in the cell-interleaved PLP data intotime units. In that case, the time interleaver 604160 uses two FECblocks for interleaving. Through this process, as pairs of cells aregenerated from two different FEC blocks, the receiver can improvereception by ensuring the diversity of the FEC blocks.

The second BICM encoding block 604200 includes a FEC encoder 604210, abit-interleaver 604220, a second demultiplexer 604230, a firstconstellation mapper 604240-1 and a second constellation mapper604240-2, and a MIMO encoder 604250, a first cell interleaver 604260-1and a second interleaver 604260-2, and a first time interleaver 604270-1and a second cell interleaver 604270-2.

The FEC encoder 604210 and the bit-interleaver 604220 can perform thesame function as the FEC encoder 604110 and the bit-interleaver 604120of the MISO method.

The second demultiplexer 604230 can transmit the PLP data bydemultiplexing to two routes necessary for MIMO transmission in additionto performing the same function as the first demultiplexer 604130 of theMISO method. In that case, the character of the data transmission foreach route may be different. Thus, the second demultiplexer can randomlyallocate the bit-interleaved PLP data into each route.

The first constellation mapper 604240-1 and the second constellationmapper 604240-2 can operate the same function as the constellationmapper 604140 of the MISO method.

The MIMO encoder 604270 performs MIMO encoding in the time-interleavedPLP data from by using MIMO encoding matrix and transmit MIMO PLP datato two routes (STx_m, STx_m+1). The MIMO encoding matrix of the presentinvention includes a spatial multiplexing, a Golden code (GC), afull-rate full diversity code, and a linear dispersion code.

The first cell interleaver 604260-1 and the second cell interleaver604260-2 can perform cell-interleaving in only a half of the PLP data inone of the FEC blocks from the routes. Thus, the first cell interleaver604260-1 and second cell interleaver 604260-2 can operate the same asthe one cell interleaver. Also, in order to execute data from aplurality of routes, as the first cell interleaver 604260-1 and thesecond cell interleaver 604260-2 are not allocated additional memory,there is an advantage of performing cell interleaving by using thememory of the one cell interleaver.

The first time interleaver 604270-1 and the second time interleaver604270-2 can operate the same as the time interleaver 604170-1, 604170-2of the MISO method. In that case, the first time interleaver 604270-1and the second time interleaver 604270-2 can be performed the same timeinterleaving or a different time interleaving.

L-signaling information includes L1-pre signaling information and L-postsignaling information. It can independently perform MISO encoding in theL1-pre signaling information and L1-post signaling information.

Thus, the third BICM encoding block 604300 includes a first encodingblock 604400 executing the L1-pre signaling information and the secondencoding block 604500 executing the L1-post signaling information.

The first encoding block 604400 includes an FEC encoder 604410, aconstellation mapper 604420, a MISO encoder 604430, cell interleavers604440-1, 604440-2, and time interleavers 604450-1, 604450-2. The secondencoding block 604500 includes a FEC encoder 604510, a bit interleaver604520, demultiplexer 604530, a constellation mapper 604540, a MISOencoder 604560, cell interleavers 604560-1, 604560-2, and timeinterleavers 604570-1, 604570-2.

The L1-pre signaling information includes information necessary fordecoding L-post signaling information and the L-post signalinginformation includes information necessary for restoring datatransmitted from the receiver.

That is, the receiver needs to decode the L1-pre signaling informationquickly and correctly for decoding the L-signaling information and thedata. Thus, the receiver of the present invention does not performbit-interleaving and de-multiplexing for the L1-pre signalinginformation in order to perform the fast decoding.

The description of first encoding block 604400 and the second encodingblock 604500 is omitted because they perform the same function as thefirst BICM block 604100.

As a result, to execute the L1-pre signaling information, the firstencoding block 604400 performs MISO encoding in the L1-pre signalinginformation and transmits the free-signaling data to two routes(STx_pre, STx_pre+1). Also, to execute L-post signaling information thesecond encoding block 604500 performs MISO encoding in the L1-postsignaling information and transmits the L1-post signaling data to tworoutes (STx_post, STx_post+1).

FIG. 16 shows a BICM encoder according to another embodiment of thepresent invention.

The BICM encoder shown in FIG. 16 according to the second embodiment isanother embodiment of the BICM encoder 101300 in FIG. 1.

The BICM encoder according to the second embodiment performsbit-interleaving in a plurality of PLP data after performinginput-processing, L1-pre signaling information, and L1-post signalinginformation, and encoding for correcting errors.

Also, the BICM encoder independently performs MISO and MIMO encoding inPLP data.

The BICM encoder in FIG. 16 includes a first BICM encoding block 607100performing MISO encoding in PLP data, a second BICM encoding block607200 performing MIMO encoding in PLP data, and a third BICM encodingblock 607300 performing MIMO encoding in signaling information.

As the BICM encoding blocks in FIG. 16 operate the same as the BICMencoding blocks in FIG. 15, further description of them is omitted.However, the BICM encoding blocks of the MISO encoder 607120, 607420,607520 and the MIMO encoder 607220 are positioned at the end of the timeinterleaver 607110, 607210-1-2, 607410 and 607510 which isdistinguishable from the BICM encoding blocks according to the firstembodiment.

Although not illustrated in FIG. 16, the BICM encoder according to thethird embodiment of the present invention may include a first BICMencoding block for processing of MISO PLP data to be MISO encoded, asecond BICM encoding block for processing of MIMO PLP data to be MIMOencoded, and a third BICM encoding block for processing of signalinginformation to be MISO encoded. The BICM encoding blocks according tothe third embodiment operate in the same way as the BICM encoding blocksaccording to the first embodiment illustrated in FIG. 15, and thus, adetailed description thereof is omitted. However, the BICM encodingblocks according to the third embodiment is different from the BICMencoding blocks according to the first embodiment in that the BICMencoding blocks according to the third embodiment do not include a MISOencoder and a MIMO encoder.

In addition, the BICM encoder according to the fourth embodiment of thepresent invention is almost the same as the BICM encoder according tothe third embodiment, except that the BICM encoder performs MIMOencoding on MIMO PLP data to be processed using the MIMO scheme. Thatis, the BICM encoder according to the fourth embodiment of the presentinvention may include a first BICM encoding block for processing MISOPLP data to be MISO encoded, a second BICM encoding block for processingof MIMO PLP data to be MIMO encoded, and a third BICM encoding block forprocessing of signaling information to be MISO encoded. Here, the thirdBICM encoding block may include a first encoding block for processing ofL1-pre signaling information and a second encoding block for processingof L1-post signaling information. In particular, the first BICM encodingblock according to the fourth embodiment may not include a MISO encoderand the second 2 BICM encoding block may include a MIMO encoder. In thiscase, the MIMO encoder may be positioned behind a time interleaver as inthe first embodiment, or may be positioned behind a constellation mapperaccording to the second embodiment as in the second embodiment. Theposition of the MIMO encoder may be changed according to a designer'sintention.

FIG. 17 shows a frame builder according to an embodiment of the presentinvention.

The frame builder shown in FIG. 17 is an embodiment of the frame builder101400 shown in FIG. 1.

The first BICM encoding block 604100 transmits MISO PLP data to tworoutes (STx_k, STx_K+1) and the second BICM encoding block 604200transmits MIMO PLP data to two routes (STx_m, STx_m+1). Also, the thirdBICM encoding block 604300 transmits the L1-pre signaling informationand the L-post signaling information to two routes (STx_pre, Stx_pre_1and STx_post, STx_post+1).

Each data is inputted into the frame builder. In that case, as shown inFIG. 17, the frame builder includes a first route receiving the BICMencoded data from STx_0 to STx_post, and a second route receiving theBICM encoded data from STx_0+1 to Stx_post+1. The data received in thefirst route is transmitted through a first antenna (Tx_1) and the datain the second route is transmitted through a second antenna (Tx_2).

As shown in FIG. 17, the frame builder according the first embodimentincludes a first frame building block 605100 executing the data from thefirst route and a second frame building block 605200 executing the datafrom the second route. The first frame building block 605100 includes afirst delay compensator 604110, a first pair-wise cell mapper 605120,and a first pair-wise frequency interleaver 605300-1, and a second framebuilding block 605200 includes a second delay compensator 605100-2executing the data from the second route, a second pair-wise cell mapper605200-2, and a second pair-wise frequency interleaver 605300-2.

The first pair-wise cell mapper 605120 and the first pair-wise frequencyinterleaver 605130, or the second pair-wise cell mapper 605120 and thesecond pair-wise frequency interleaver 605310 operate independently butthe same functions in the first and the second routes respectively.

A method of performing data in the first frame building block 605100 andthe second frame building block 605200.

The first delay compensator 605110 and the second delay compensator605110 can compensate the L1-pre signaling data or the L1-post signalingdata for the delay in the first frame and by the BICM encoder 604300.The L-signaling information can include information not only in thecurrent frame but also in the next frame. Thus, during the inputprocessing, the L1-signaling information can be delayed one frame asopposed to PLP data inputted in the current frame. Through this process,one frame of the L1-signaling information having information about thecurrent and the next frames.

The first pair-wise cell mapper 605120 and the second pair-wise cellmapper 605220 can perform mapping in the PLP data and the L1-signalingdata in symbol units into cell units in a frame in the sub-carrier ofthe OFDM symbols.

In that case, the PLP data includes a common PLP DATA, a MISO/MIMOencoded PLP data and a sub-slice processor module 605120-1-2 performsframe-mapping in the PLP data in cell units for the diversity effect.

Also, the first pair-wise cell mapper 605120 and the second pairwaisecell mapper 605220 can perform frame-mapping in two consecutive inputtedcells in pairs.

For the better restoration performance of MISO signals, coherencebetween MSI transmitting channels should be secured when performing MISOencoding. Thus, in order to secure coherence, the first pair-wise cellmapper 605120 and the second pair-wise cell mapper 605220 pair up cellsgenerated from the same PLP and perform OFDM modulating in the paired-upcells. Then coherence between the channels will be maximized. In otherwords, according to an embodiment of the present invention, as the MISOencoder is positioned in the front of the BICM encoder, the structure ofthe frames is in pairs considering such MISO encoding process.

As mentioned above, when performing bit-interleaving or timeinterleaving by the bit-interleaver 604120 and the time interleaver604160 using two FEC blocks, two paired up cells can be generated fromtwo different FEC blocks. As the receiver ensures diversity, higherreception can be obtained. The first pair-wise frequency interleaver605130 and the second pair-wise frequency interleaver 605230 performfrequency interleaving in the data in cell units from each route andtransmits the frequency-interleaved data to the OFDM generator througheach route.

In that case, the first pair-wise frequency interleaver 605130 and thesecond pair-wise frequency interleaver 605230 pair up two consecutivecells in interleaving units and then perform frequency interleaving.This is to maximize coherence between channels.

The frame builder illustrated in FIG. 17 may be applied to the first andsecond embodiments of the present invention. According to the third andfourth embodiments of the present invention, the frame builder mayinclude a first cell mapper and a second cell mapper instead of thefirst pair-wise cell mapper 605120 and the second pair-wise cell mapper605220, and include a first frequency interleaver and a second frequencyinterleaver instead of the first pair-wise frequency interleaver 605130and the second pair-wise frequency interleaver 605230.

According to the third embodiment, after frequency interleaving, thatis, after MISO/MIMO encoding in the OFDM generating process, MIMO/MISOencoding can be done in OFDM symbol units. If the MISO PLP data cellsand MIMO PLP data cells are mapped in the same OFDM symbol, the OFDMgenerator cannot perform MISO/MIMO encoding independently. Thus, thefirst cell mapper and the second cell mapper dose not map the MISO/MIMOPLP data in the same OFDM symbol.

Also, in order to simplify the transmitting system, the first and secondcell mappers according to the third embodiment operate the same.

But, although the MISO PLP data, L1-pre and post signaling data istransmitted from the first route only, the MIMO PLP data can be from thefirst and the second routes. Therefore, depending on the data inputted,the performance of the cell mapper is different.

More detailed description is as follows.

First, the first cell mapper and the second cell mapper receive the sameMISO PLP data from the first route and the same L1-pre and postsignaling data from the delay compensator. In that case, the first cellmapper and the second cell mapper perform mapping in the inputted datato be allocated into a sub-carrier of the OFDM symbol.

Second, among the first cell mapper and the second cell mapper, thefirst cell mapper only receives the MISO PLP data and the delayedcompensated L1-pre and post signaling data. In that case, the secondcell mapper performs mapping only for the MIMO PLP.

The first frequency interleaver and the second frequency interleaverperform frequency interleaving in the inputted data by cell units andtransmits the data to the OFDM generator.

In that case, the first frequency interleaver and the second frequencyinterleaver perform frequency interleaving in the OFDM symbol intointerleaving units. Also, if the second cell mapper 619210 receives MIMOPLP data only, the second frequency interleaver also performsinterleaving in MIMO PLP data only.

FIG. 18 shows an OFDM generator according to an embodiment of thepresent invention.

The OFDM generator in FIG. 18 is an embodiment of the OFDM generator101500 shown in FIG. 1.

The present invention transmits broadcast signals by the MISO/MIMOmethod through two antennas. The OFDM generator in FIG. 19 receives anddemodulates the broadcast signals through a first and a second route. Itthen transmits the signals to two antennas (Tx1, Tx2).

A first OFDM generating block 606800 modulates the broadcast signalsthrough the first antenna (Tx1) and a second OFDM generating block606900 modulates the broadcast signals through the second antenna (TX2).

If channel correlation between the first and second antennas is large,transmitted signals can apply polarity depending on the channelcorrelation. In the present invention, such a method is called polaritymultiplexing MIMO. The first antenna is called “vertical antenna” andthe second antenna is called “horizontal antenna”. The first OFDMgenerating block 606800 performs OFDM modulating in broadcast signalsthrough the first antenna (Tx1) and the second transmitter 606900performs OFDM modulating in the broadcast signals from the first routeand transmits the signals to the second antenna (Tx2).

Modules including the first OFDM generating block 606800 and the secondOFDM generating block 606900 are as follows.

The first OFDM generating block 606800 includes a pilot insertion module606100-1, an IFFT module 606200-1, a PAPR module 606300-1, a GIinsertion module 606400-1, a P1 symbol insertion module 606500-1, an AP1symbol insertion module 606600-1 and a DAC 606700-1, wherein modules inthe first transmitting unit 606800 operate the same functions.

The second OFDM generating block 606900 includes a pilot insertionmodule 606100-2, an IFFT module 606200-2, a PAPR module 606300-2, a GIinsertion module 606400-2, a P1 symbol insertion module 606500-2, an AP1symbol insertion module 606600-2 and a DAC 606700-2, wherein modules inthe first transmitting unit 606800 operate the same functions.

Thus, modules in the first OFDM generating block 606800 will beillustrated in more detail. The pilot insertion module inserts a pilotof the predetermined pilot pattern into a frame and transmits it to theIFFT module 606200-1. The pilot pattern information is transmitted withAP1 signaling information or L1-signaling information.

The IFFT module 606200-1 performs IFFT algorithm in the signals andtransmits them to the PAPR module 606300-1.

The PAPR module 606300-1 reduces PAPR of the signals in a time domainand transmits them to the GI insertion module 606400-1. Also, feedbackon necessary information based on the PAPR reduction algorithm is givento the pilot insertion module 606100-1.

The GI insertion module 606400-1 copies the end of the effective OFDMsymbol, inserts guard intervals in cyclic prefix to each OFDM symbol,and transmits them to the P1 symbol insertion module 606500-1. The GIinformation can be transmitted through the P1 signaling information orL1-pre signaling information.

The P1 and AP1 symbol are inserted in every frame of the P1 insertionmodule in the OFDM generator. That is, the P1 insertion module caninsert more than two preamble symbols in every frame. When using morethan two preamble symbols, burst fading that can happen in the mobilefading conditions will be more strengthened and signal detectionperformance will be improved.

The P1 symbol insertion module 606500-1 inserts a P1 symbol in thebeginning of each frame and transmits it to the AP1 symbol insertionmodule 606600-1.

The AP1 symbol insertion module 606600-1 inserts an AP1 symbol at theend of the P1 symbol and transmits it to the DAC 606700-1.

The DAC 606700-1 converts the signal frame having the P1 symbol to ananalog signal and transmits it to the transmitting antenna (Tx1).

The OFDM generator shown in FIG. 18 may be applied to the first andsecond embodiments of the present invention.

Although not shown in FIG. 18, according to the third embodiment of thepresent invention, the OFDM generator may include a MISO/MIMO encoder, afirst OFDM generating block, and a second OFDM generating block. Thefirst OFDM generating block and the second generating block according tothe third embodiment of the present invention may perform the sameoperations as those of the first OFDM generating block 606800 and thesecond OFDM generating block 606900.

If the input data is MISO PLP data or L1-pre and post signaling datafrom the first and second routes, the MIMO/MISO encoder 603100 performsMISO encoding in the data into OFDM symbol units by using MISO encodingmatrix and transmits it to the first and second generating blocks620200, 620300. In that case, the input data is transmitted from eitherof the first or second route. According to an embodiment, the MISOencoding matrix can include an OSTBC (Orthogonal Space-Time BlockCode)/OSFBC(Orthogonal Space Frequency Block Code/Alamouti code).

If data from the first and second routes is MIMO PLP data, the MIMO/MISOencoder performs MIMO encoding in the data into OFDM symbol units byusing MIMO encoding matrix and transmits it to the first and second OFDMgenerating blocks. The MIMO encoding matrix of the present inventionincludes a spatial multiplexing, a Golden code (GC), a full-rate fulldiversity code, and a linear dispersion code. Also, the MIMO encoderperforms MIMO encoding by using MIMO encoding matrix.

In addition, the OFDM generator according to the fourth embodiment ofthe present invention may include a MISO encoder, a first OFDMgenerating block, and a second OFDM generating block. The first OFDMgenerating block and the second generating block according to the fourthembodiment of the present invention may perform the same operations asthose of the first OFDM generating block 606800 and the second OFDMgenerating block 606900.

The MISO encoder performs MISO encoding in the frequency-interleavedMISO PLP data, L1-pre signaling data and L-post signaling data. The MISOencoder operates the same as the MIMO/MISO encoder according to thethird embodiment. In addition, if the MIMO encoded MIMO PLP data isinputted, it may be bypassed and the MISO encoder may perform MISOencoding in the MIMO encoded MIMO PLP data.

FIGS. 19 to 23 show a structure block of the broadcast signal receiveraccording to an embodiment of the present invention.

FIG. 19 shows an OFDM demodulator according to an embodiment of thepresent invention.

FIG. 19 shows a drawing of the OFDM demodulator according to anembodiment of the OFDM demodulator 107100 illustrated in FIG. 2.

According to an embodiment of the present invention, the presentinvention requires two antennas, Rx1 and Rx2, to receive transmittedsignals by MIMO/MISO. The OFDM demodulator shown in FIG. 20 can performOFDM demodulation through the Rx1 and Rx2 antennas.

A block demodulating transmitted signals through a first antenna (Rx1)is called a first OFDM demodulating block 610100 and a blockdemodulating transmitted signals through a second antenna (Rx2) iscalled a second OFDM demodulating block 610200.

In addition, the present invention can utilize polarity multiplexingMIMO according to an embodiment of the present invention. The first OFDMdemodulating block 610100 performs OFDM demodulation in the broadcastsignals transmitted through the first antenna (Rx1) and outputs thesignals by a frame demapper to a first route, and the seconddemodulating block 610200 performs OFDM demodulating in the broadcastsignals transmitted through the second antenna (Rx2) and outputs thesignals by a frame demapper to a second route.

Also, the OFDM according to the first embodiment in FIG. 19 can performthe reverse process of the OFDM generator illustrated in FIG. 18.

The first OFDM demodulating block 610100 and the second OFDMdemodulating block 610200 included in OFDM demodulator according to anembodiment of the present in invention including are as follows.

The first OFDM demodulating block 610100 includes a tuner 610110, an ADC610120, a P1 symbol detection module 610130, an AP1 symbol detectionmodule 610140, a synchronizing module 610150, a GI cancellation module610160, a FFT module 610170 and a channel estimation module 610180.

The second OFDM demodulating block 610200 comprises a tuner 610210, anADC 610220, a P1 symbol detection module 610230, an AP1 symbol detectionmodule 610240, a synchronizing module 610250, a GI cancellation module610260, a FFT module 610270 and a channel detection module 610280, andoperates the same as the first OFDM demodulating block 610100.

Thus, modules in the first OFDM demodulating block 610100 will befurther illustrated.

The tuner 610110 receives broadcast signals by selecting a frequencyrange and transmits it to the ADC 610120 by compensating the size of thesignal.

The ADC 610120 coverts analog broadcast signals into digital signals andtransmits them to the P1 symbol detection module 610130.

The P1 symbol detection module 610130 extracts P1 symbols in the P1signaling information and decodes the P1 signaling information. Also,the P1 symbol detection module 610130 transmits the decoded P1 signalinginformation to the synchronizing module 610150 and a system controller(not shown in the drawing). The system controller determines which framethe received signal has by using the decoded P1 signaling informationand controls other devices.

The AP1 symbol detection module 610140 extracts AP1 symbols in the AP1signaling information and decodes the AP1 signaling information. Also,the AP1 symbol detection module 610140 transmits the decoded AP1signaling information to the synchronizing module 610150 and a systemcontroller (not shown in the drawing). The system controller determinesthe pilot pattern information in the current frame and L1-pre spreadinterval information by using the decoded AP1 signaling information.

The synchronizing module 610150 performs time and frequencysynchronizing by using the decoded P1 signaling information and the AP1signaling information.

The GI cancellation module 610160 deletes guard intervals included inthe synchronized signals and transmits them to the FFT module 610170.

The FFT module 610170 converts the signals from the time domain to thefrequency domain by performing FFT algorithm.

The channel detection module 610180 detects a transmitting channel fromthe transmitting antenna to the receiving antenna by using pilot signalshaving the converted signals. Then, the channel detection module 610180can additionally perform equalizing for each of the received data.Signals that are converted into the frequency domain will be inputted inthe frame demapper.

The OFDM demodulator illustrated in FIG. 19 may be applied to the firstand second embodiments of the present invention.

Although not illustrated in FIG. 19, according to the third embodimentof the present invention, the OFDM demodulator may include a first OFDMdemodulating block, a second OFDM demodulating block, and a MISO/MIMOdecoder. The first OFDM demodulating block and the second OFDMdemodulating block according to the third embodiment of the presentinvention may perform the same operations as those of the first OFDMdemodulating block 610100 and the second OFDM demodulating block 610200.However, the OFDM demodulator according to the third embodiment mayinclude a MIMO/MISO decoder 626300, a detailed operation of which willbe described below.

The OFDM according to the fourth embodiment of the present invention mayinclude a first OFDM demodulating block, a second OFDM demodulatingblock, and a MISO decoder. The first OFDM demodulating block and secondOFDM demodulating block according to the fourth embodiment of thepresent invention may perform the same operations as those of the firstOFDM demodulating block 610100 and the second OFDM demodulating block610200.

FIG. 20 shows a frame demapper according to an embodiment of the presentinvention.

The frame demapper in FIG. 20 is an embodiment of the frame demapper107200 in FIG. 2.

The frame demapper illustrated in FIG. 20 includes the first framedemapping block 611100 executing data from a first route and a secondframe demapping block 611200 executing data from a second route. Thefirst frame demapping block 611100 includes a first pair-wise frequencydeinterleaver 611110 and a first pair-wise cell demapper 611120, and thesecond demapping block 611200 includes a second pair-wise frequencydeinterleaver 611210 and a second pair-wise cell demapper 611220.

Also, the first pair-wise frequency deinterleaver 61110 and the firstpair-wise cell demapper 611120 or the second pair-wise frequencydeinterleaver 611210 and the second pair-wise cell demapper 611220 canoperate independently and the same through a first route and a secondroute respectively.

Also, the frame demapper illustrated in FIG. 20 can perform the reverseprocess of the frame builder illustrated in FIG. 17.

A method of performing data by blocks included in the first framebuilder demapping block 611100 and the second frame builder demappingblock 611200 is as follows.

The first pair-wise frequency deinterleaver 611110 and the secondpair-wise frequency deinterleaver 611210 perform deinterleaving in datain the frequency domain through the first and second routes into cellunits in that case, the first pair-wise frequency deinterleaver 611110and the second frequency deinterleaver 611210 pair up two consecutivecells in deinterleaving units and perform frequency deinterleaving. Thedeinterleaving process can be performed in a reverse direction of theinterleaving process in the transmitting unit. Thefrequency-deinterleaved data will be transmitted in the original order.

The first pair-wise cell demapper 611120 and the second pair-wise celldemapper 611220 can extract common PLP data, PLP data and L-signalinginformation in cell units from the de-interleaved data. The extractedPLP data includes MISO PLP data for the MISO method and MIMO PLP datafor the MIMO method, and the extracted L1-signaling data includesinformation necessary for the current and next frames. Also, if thetransmitter performs sub-slicing in the PLP data, the first and thesecond pair-wise cell demappers 611120, 611220 can merge the sliced PLPdata and generate it in one stream.

Also, the first pair-wise cell demapper 611120 and the second pair-wisecell demapper 611220 can pair up two consecutive cells.

Data transmitted through the first route is inputted to the BICM decoderby the route from SRx_0 to SRx_post and data transmitted through thesecond route is inputted to the BICM decoder by the route from SRx_0+1to SRx_post+1.

The frame demapper shown in FIG. 20 may be applied to the first andsecond embodiments of the present invention. In accordance with thethird and fourth embodiments of the present invention, the framedemapper includes a first frame demapping block 627100 performing datafrom a first route and a second frame demapping block 627200 performingdata from a second route.

The first frame demapping block includes a first frequencydeinterleaver, a first cell demapper, a first combiner, a secondcombiner and a third combiner, and the second frame demapping blockincludes a second frequency deinterleaver and a second cell demapper.

Also, the first frequency deinterleaver and the first cell demapper orthe second frequency deinterleaver and the second cell demapper canoperate independently and the same through a first route and a secondroute respectively.

The first frequency deinterleaver and the second frequency deinterleaverperform deinterleaving in data in the frequency domain through the firstand second routes into cell units.

The first and second cell demappers perform extracting common PLP data,PLP data and L-signaling data from the deinterleaved data by cell units.The extracted PLP data includes the MISO decoded MISO PLP data and MIMOdecoded MIMO PLP data, and the extracted L1-signaling data includesinformation necessary for the current and next frames. Also, if thetransmitter performs sub-slicing in the PLP data, the first sub-sliceprocessor 627120-1, 627220-1 of the first and the second cell demappers627120, 627220 can merge the sliced PLP data and generate it in onestream.

The first combiner can combine the MISO decoded MISO PLP data if it doesnot combine the MISO PLP data in the MIMO/MISO decoder.

The second combiner and the third combiner can operate the same as thefirst combiner but it deals with L1-pre and L1-post signaling data.

FIG. 21 shows a BICM encoder according to an embodiment of the presentinvention.

The BICM encoder in FIG. 21 according to the first embodiment of thepresent invention is an embodiment of the BICM encoder 107300 in FIG. 2.

The BICM decoder according to the first embodiment receives data fromthe first route via SRx_0 to SRx_post by a frame demapper and data fromthe second route via SRx_0+1 to SRx_post+1 and performs BICM decoding.

Also, the BICM decoder according to the first embodiment independentlyperforms MISO/MIMO encoding in each of the data.

That is, the BICM decoder in FIG. 21 includes a first BICM decodingblock 612100 performing MISO PLP data from SRx_k and SRx_k+1, a secondBICM decoding block 612200 performing MIMO PLP data from SRx_m andSRx_m+1, and a third BICM decoding block 612300 performing MISO encodingin the L-signaling information from SRx_pre, SRx_pre+1, SRx_post, andSRx_post+1.

Also, the BICM decoder according to the first embodiment of the presentinvention can perform the reverse process of the BICM encoder accordingto the first embodiment of the present invention illustrated in FIG. 15.

Data-preformation method for each block is illustrated.

First, the first BICM decoding block 612100 includes a timedeinterleaver 612110-1, 612110-2, a cell deinterleaver 612120-1,612120-2, a MISO decoder 612130, a constellation demapper 612140, afirst demultiplexer 612150, a bit deinterleaver 612160, and a FECdecoder 612170.

The time deinterleaver 612110-1, 612110-2 restores the MISO decoded datainto a time domain and the cell deinterleaver 612120-1, 612120-2performs deinterleaving in the time-deinterleaved data into cell units.

The MISO decoder 612130 can perform MISO decoding in MISO PLP data.

The MISO decoder 612130 can perform four functions.

First, if the channel estimation modules 610180, 610280 included in theOFDM demodulator illustrated in FIG. 19 do not perform channelequalizing, the MISO decoder 612130 applies the effect of the channeldetection regarding every transmissible reference point and computes anLLR value. Therefore, it will have the same effect.

Second, the MISO decoder 612130 performs the following functions basedon the performance of the constellation mapper 604140. If the BICMencoder of the broadcast signal transmitter rotates the constellationmapper with a certain angle and delays the Q-phase element of theconstellation for a certain value, the MISO decoder 612130 delays theI-phase element of the constellation for a certain value and computes a2D-LLR value based on the rotation angle.

If the constellation mapper 604140 does not rotate constellation anddoes not delay the Q-phase of constellation for a certain value, theMSIO decoder 612130 can compute the 2-D LLR value based on the normalQAM.

Third, the MISO decoder 612130 selects a decoding matrix to perform thereverse process based on the encoding matrix used by the MISO encoder604150.

Fourth, the MISO decoder 612130 can combine signals inputted from twoantennas. The signal combining method includes maximum ratio combining,equal gain combining, and selective combining and obtains the diversityeffect by maximizing the SNR of the combined signals.

The MISO decoder 612130 performs MISO decoding in the combined signaland combine the MISO-decoded combined signals.

The constellation demapper 612140 can perform the following functionsbased on the performance of the MISO decoder 612130.

First, if the MISO decoder 612130 does not transmit the LLR valuedirectly and only performs MISO decoding, the constellation demapper612140 can compute the LLR value. In more detail, if the constellationdemapper 604140 in the BICM encoder performs constellation rotation orQ-phase element delay, the constellation demapper 612140 delay theI-phase LLR element and computes the LLR value. If the constellationdemapper 604140 does not perform the constellation rotation and Q-phaseelement delay, the constellation demapper 612140 can compute the LLRvalue based on the normal QAM.

The computing the LLR value includes computing 2-D LLR and computing 1-DLLR. When computing the 1-D LLR, the complexity of the LLR computationcan be reduced by executing either one of a first or a second route.

The first multiplexer 612150 restores demapped data in bit stream.

The bit-interleaver 612160 performs deinterleaving in the bit-stream,FEC decoding in the deinterleaved data, and outputs MISO PLP data bycorrecting errors in the transmitting channels.

The second BICM decoding block 612200 includes a first timedeinterleaver 612210-0 and a second time deinterleaver 612210-1, a firstcell deinterleaver 612220-0 and a second cell deinterleaver 612220-1, aMIMO decoder 612230, a first constellation demapper 612240-0 and asecond constellation demapper 612240-1, a second multiplexer 612250, abit interleaver 612260 and a FEC decoder 612270.

The first time deinterleaver 612210-0 and the second time deinterleaver612210-1 perform deinterleaving in the MIMO decoded data into cellunits. In that case, the first cell deinterleaver 612220-0 and thesecond deinterleaver 612220-1 performs cell deinterleaving in only ahalf of the cell data in one FEC block. As a result, cell deinterleavingby the first and second cell deinterleaver 612220-0, 612220-1 has thesame effect as deinterleaving by a cell deinterleaver using one FECblock.

The MIMO decoder 612230 performs in MIMO PLP data from SRx_m andSRx_m+1. The MIMO decoder 612210 can perform the four functions of theMISO decoder 612110 except for the fourth function in which the signalsare to be combined. Then, the MIMO decoder 612230 performs decoding byusing MIMO encoding matrix of the first and sixth embodiment.

The first constellation demapper 612240-0, the second constellationdemapper 612240-1, the second multiplexer 612250, bitinterleaver 612260and FEC decoder 612270 operates the same as those according to the firstBICM decoding block 612100.

The third BICM decoding block 612300 includes a first decoding block612400 performing L1-pre signaling data and a second decoding block612500 performing L1-post signaling data. The first decoding block612400 includes a time deinterleaver 612410-1, 612410-2, a celldeinterleaver 612420-1, 612420-2, a MISO decoder 612430, a constellationdemapper 612440, and a FEC decoder 612450, and the second decoding block612500 includes a time deinterleaver 612510-1, 612510-2, a celldeinterleaver 612520-1, 612520-2, a MISO decoder 612530, a constellationdemapper 612540, a multiplexer 612550, a bit deinterleaver 612560, and aFEC decoder 612570.

As the first decoding block 612400 and the second decoding block 612500have the same functions, the description of the first BICM decodingblock 612100 is omitted.

As a result, the first BICM decoding block 612100 outputs the BICMdecoded MISO PLP data to an output processor and the second BICMdecoding block 612200 transmits the BICM decoded MIMO PLP data to theoutput processor.

The first decoding block 612400 in the third BICM decoding block 612300performs MSIO decoding in L1-pre signaling data and transmits the data.Also, the second decoding block 612500 in the third BCIM decoding block612300 performs MISO decoding in L1-post signaling data and transmitsone L1-post signaling information.

FIG. 22 shows a BICM decoder according to another embodiment of thepresent invention.

The BICM decoder in FIG. 22 according to the second embodiment of thepresent invention is an embodiment of the BICM decoder 107300 in FIG. 2.

The BICM decoder according to the second embodiment receives datatransmitted from a first route to a route of from SRx_0 to SRx_post anddata transmitted from a second route to a route of from SRx_0+1 toSRx_post+1, and performs BICM decoding. Also, the BICM decoder accordingto the second embodiment can independently apply the MISO/MIMO process.

That is, the BICM decoder in FIG. 22 includes a first BICM decodingblock 615100 performing MISO encoding in MSIO PLP data from SRx_k andSRx_k+1, a second BICM decoding block 615200 performing in MIMO PLP datafrom SRx_post and SRx_post+1, and a third BICM decoding block 615300performing MISO encoding in L1-signaling data from SRx_pre, SRx_pre+1,SRx_post, and SRx_m+1.

Also, the third BICM decoding block 615300 includes a first decodingblock 615400 performing the L1-pre signaling data and a second decodingblock 615500 performing L1-post signaling data.

Also, the BICM decoder according to the second embodiment can performthe reverse process of the BICM encoder according to the secondembodiment illustrated in FIG. 16.

The decoding blocks according to the second embodiment in FIG. 22operate the same as the decoding blocks according to the firstembodiment in FIG. 21. Therefore, further description is omitted.However, the only difference is that in the BICM decoder the MISOdecoder 615110, 615410, 615510 and the MIMO decoder 615310 are locatedin front of the time deinterleaver deinterleaver 615120, 615220-1,615220-2, 615420, 615520.

As above described, the PLP data and the signaling information areperformed into symbol units after the constellation mapping. Inaddition, the broadcast signal receiver may perform BICM decoding ondata received in reverse processes to those of the BICM encoding blocksaccording to the first embodiment or the second embodiment. In thiscase, a MISO decoder, a MIMO decoder, a time deinterleaver, and a celldeinterleaver of the broadcast signal receiver may perform the receiveddata in symbol units. However, the BICM decoder of the broadcast signalreceiver may first perform MISO decoding or MIMO decoding for each data,and thus, each data is output in bit units. Then, the BICM decoder ofthe broadcast signal receiver may perform time deinterleaving and celldeinterleaving processes, but requires information regarding a symbolunit of data output in bit units. Thus, the broadcast signal receivermay store information regarding symbol mapping of input bits requiredfor the deinterleaving processes.

As not shown in drawings, the BICM decoder according to the thirdembodiment includes a first BICM decoding block processing the MISOdecoded MISO PLP data transmitted through one path, a second BICMdecoding block processing the MIMO decoded MIMO PLP data transmittedthrough two paths, and a third BICM decoding block processing the MISOencoded L-signaling data transmitted through two paths. Also, the thirdBICM decoding block includes a first decoding block processing L1-presignaling data and a second decoding block processing L1-post signalingdata.

Also, as the BICM decoder according to the third embodiment operates thesame as the BICM encoding blocks according to the first embodiment inFIG. 21. However, the only difference is that the BICM decoding blocksaccording to the third embodiment do not include MISO/MIMO decoders.

Also, the BICM decoder according to the fourth embodiment of the presentinvention includes a first BICM decoding block processing MISO PLP datatransmitted through one path, a second BICM decoding block processingMIMO PLP data transmitted through two paths, and a third BICM decodingblock processing MISO decoded L1-signaling data transmitted through twopaths.

Also, the third BICM decoding block includes a first decoding blockprocessing L-signaling data and a second decoding block processingL-post signaling data.

As the first BICM decoding block according to the fourth embodimentoperates the same as the BICM decoding blocks illustrated in FIG. 21.

But, the only difference is that the second BICM decoding block includesthe MIMO decoder as opposed to the third embodiment of the presentinvention. In that case, the transmitting character of MIMO PLP datafrom a first and a second route may or may not be the same. Also, if themodulation orders of the two MIMO PLP data are the same, a secondconstellation mapper, a second cell interleaver and a second timeinterleaver may not be used. Thus, two of the MIMO PLP data will bemerged into one input in the first time deinterleaver, the first celldeinterleaver, the first constellation demapper, and then will beinputted to the second multiplexer. In addition, the MIMO decoder may bepositioned in front of the time deinterleavers as in the firstembodiment or may be positioned in front of the constellation demappersas in the second embodiment.

FIG. 23 and FIG. 24 show an embodiment of an output processor includedbroadcast signal receiver according to an embodiment of the presentinvention. The following is a specific description of the outputprocessor.

FIG. 23 shows an output processor of the broadcast signal receiveraccording to an embodiment.

The output processor in FIG. 23 is an embodiment of the output processor107400 in FIG. 2.

The output processor in FIG. 23 as opposed to an input processorperforming single PLP in FIG. 12 performs the reverse process of it andincludes a BB discrambler 616100, a padding remove module 616200, aCRC-8 decoder 616300 and a BB frame processor 616400. The outputprocessor performs the reverse process of the input processorillustrated in FIG. 13 by receiving bit stream from the BICM decoder.

The BB descrambler 616100 receives bit stream, performs XOR algorithmwith the same bit-string as PRBS processed by the BB scrambler andoutputs it. The padding remove module 616200 removes, if necessary,padding bits inserted in the padding insertion module. The CRC-8 decoder616300 performs CRC decoding in the bit-stream and the BB frameprocessor 616400 decodes information in the BB frame header and restoresTS or GS by using the decoded information.

FIG. 24 shows another embodiment of an output processor of the presentinvention.

The output processor in FIG. 24 as opposed to the input processor inFIG. 13 and FIG. 14 performing a plurality of PLP performs the reverseprocess of it. The output processor includes a plurality of blocks for aplurality of PLP. The blocks are as follows. The output processorincludes a BB descrambler 617100, 617400-1, 617400-2 and a paddingremoval module 617120, a CRC-8 decoder 617130, a BB frame processor617140, a De-jitter buffer 617150, a null packet insertion module617160, a TS clock regeneration module 617170, an in-band signalingdecoder 617180, a TS recombination module 617300 and a L1 signalingdecoder 617410. The same blocks as in FIG. 23 are omitted.

Processing for a plurality of PLP can be shown as decoding PLP dataregarding common PLP or decoding service components like scalable videoservice or a plurality of services at once. The BB descrambler 617110,the padding removal module 617120, the CRC-8 decoder 617130 and the BBframe processor 617140 operate the same as those in FIG. 23.

The De-jitter buffer 617150 compensates a temporarily inserted delay forthe synchronization of a plurality of PLP based on Time To Output (TTO)parameters. The null packet insertion module 617160 restores the deletednull packet based on the Deleted Null Packet (DNP) information. The TSclock regeneration module restores the detailed time synchronization ofthe outputted packet based on Input Stream Time Reference information.The TS recombination module 617300 receives the restored common PLP andrelated PLP data and transmit the original TS, IP or GS. The TTOparameters, DNP information, and ICSR information are obtained by the BBframe processor and it can transmit the data to each block or a systemcontroller.

The in-band signaling decoder 617200 restores in-band signalinginformation via the padding bit filed of PLP data and transmits it.

As for L1 signaling information, the BB descramblers 617400-1, 617400-2performs descrambling in the corresponding L1 pre signaling informationdata and L1-post signaling information, and the L1 signaling decoder6174100 decodes the descrambled data and restores the L1 signalinginformation. The restored L-signaling information includes L1-presignaling information and L1-post signaling information. It will also betransmitted to the system controller and provides parameters for BICMdecoding, frame demapping, and OFDM demodulating. The L1 signalinginformation can be inputted as one BB descrambler and will bedescrambled.

As described above, the Spatial Multiplexing (SM) scheme and the GoldenCode (GC) may be used as the MIMO scheme. A detailed description thereofwill be given below

FIG. 25 illustrates MIMO transmission and reception systems according toan embodiment of the present invention.

As shown in FIG. 25, the MIMO transmission system includes an inputsignal generator 201010, a MIMO encoder 201020, a first transmit antenna201030, and a second transmit antenna 201040. In the following, theinput signal generator 201010 may be referred to as a divider and theMIMO encoder 201020 may be referred to as a MIMO processor.

The MIMO reception system may include a first receive antenna 201050, asecond receive antenna 201060, a MIMO decoder 201070, and an outputsignal generator 201080. In the following, the output signal generator201080 may be referred to as a merger and the MIMO decoder 101070 may bereferred to as an ML detector.

In the MIMO transmission system, the input signal generator 201010generates a plurality of input signals for transmission through aplurality of antennas. In the following, the input signal generator201010 may be referred to as a divider. Specifically, the input signalgenerator 201010 may divide an input signal for transmission into 2input signals and output the first input signal S1 and the second inputsignal S2 for MIMO transmission.

The MIMO encoder 201020 may perform MIMO encoding on the plurality ofinput signals S1 and S2 and output a first transmission signal St1 and asecond transmission signal St2 for MIMO transmission and the outputtransmission signals may be transmitted through a first antenna 201030and a second antenna 201040 via required signal processing andmodulation procedures. The MIMO encoding 201020 may perform encoding ona per symbol basis. The SM scheme or the GC scheme may be used as theMIMO encoding method. In the following, the MIMO encoder may be referredto as a MIMO processor. Specifically, the MIMO encoder may process aplurality of input signals according to a MIMO matrix and a parametervalue of the MIMO matrix which are described below.

The input signal generator 201010 is an element that outputs a pluralityof input signals for MIMO encoding and may also be an element such as ademultiplexer or a frame builder depending on the transmission system.The input signal generator 201010 may also be included in the MIMOencoder 201020 such that the MIMO encoder 201020 generates a pluralityof input signals and performs encoding on the plurality of inputsignals. The MIMO encoder 201020 may be a device that performs MIMOencoding or MIMO processing on a plurality of signals and outputs theencoded or processed signals so as to acquire diversity gain andmultiplexing gain of the transmission system.

Since signal processing should be performed on a plurality of inputsignals after the input signal generator 201010, a plurality of devicesmay be provided next to the input signal generator 201010 to processsignals in parallel or one device including one memory may be providedto sequentially process signals or to simultaneously process signals inparallel.

The MIMO reception system receives a first reception signal Sr1 and asecond reception signal Sr2 using a first receive antenna 201050 and asecond receive antenna 201060. The MIMO decoder 201070 then processesthe first reception signal and the second reception signal and outputs afirst output signal and a second output signal. The MIMO decoder 201070processes the first reception signal and the second reception signalaccording to the MIMO encoding method used by the MIMO encoder 201020.As an ML detector, the MIMO decoder 201070 outputs a first output signaland a second output signal using information regarding the channelenvironment, reception signals, and the MIMO matrix used by the MIMOencoder in the transmission system. In an embodiment, when ML detectionis performed, the first output signal and the second output signal mayinclude probability information of bits rather than bit values and mayalso be converted into bit values through FEC decoding.

The MIMO decoder of the MIMO reception system processes the firstreception signal and the second reception signal according to the QAMtype of the first input signal and the second input signal processed inthe MIMO transmission system. Since the first reception signal and thesecond reception signal received by the MIMO reception system aresignals that have been transmitted after being generated by performingMIMO encoding on the first input signal and the second input signal ofthe same QAM type or different QAM types, the MIMO reception system maydetermine a combination of QAM types of the reception signals to performMIMO decoding on the reception signals. Accordingly, the MIMOtransmission system may transmit information identifying the QAM type ofeach transmission signal in the transmission signal and the QAM typeidentification information may be included in a preamble portion of thetransmission signal. The MIMO reception system may determine thecombination of the QAM types of the reception signals from the QAM typeidentification information of the transmission signals and perform MIMOdecoding on the reception signals based on the determination.

Thereafter, the output signal generator 201080 may generate an outputsignal by merging the first output signal and the second output signal.

The following is a description of a MIMO encoder and a MIMO encodingmethod that have low system complexity, high data transmissionefficiency, and high signal reconstruction (or restoration) performancein various channel environments according to an embodiment of thepresent invention.

The SM scheme is a method in which data is simultaneously transmittedthrough a plurality of antennas without MIMO encoding. In this case, thereceiver can acquire information from data that is simultaneouslyreceived through a plurality of receive antennas. The SM scheme has anadvantage in that the complexity of a Maximum Likelihood (ML) decoderthat the receiver uses to perform signal reconstruction (or restoration)is relatively low since the decoder only needs to check a combination ofreceived signals. However, the SM scheme has a disadvantage in thattransmit diversity cannot be achieved at the transmitting side. In thecase of the SM scheme, the MIMO encoder bypasses a plurality of inputsignals. In the following, such a bypass process may be referred to asMIMO encoding.

The GC scheme is a method in which data is transmitted through aplurality of antennas after the data is encoded according to apredetermined rule (for example, according to an encoding method usinggolden code). When the number of the antennas is 2, transmit diversityis acquired at the transmitting side since encoding is performed using a2×2 matrix. However, there is a disadvantage in that the complexity ofthe ML decoder of the receiver is high since the ML decoder needs tocheck 4 signal combinations.

The GC scheme has an advantage in that it is possible to perform morerobust communication than using the SM scheme since transmit diversityis achieved. However, such a comparison has been made when only the GCscheme and the SM scheme are used for data processing for datatransmission and, if data is transmitted using additional data coding(which may also be referred to as outer coding), transmit diversity ofthe GC scheme may fail to yield additional gain. This failure easilyoccurs especially when such outer coding has a large minimum Hammingdistance. For example, the transmit diversity of the GC scheme may failto yield additional gain compared to the SM scheme when data istransmitted after being encoded by adding redundancy for errorcorrection using a Low Density Parity Check (LDPC) code having a largeminimum Hamming distance. In this case, it may be advantageous for thebroadcast system to use the SM scheme having low complexity.

FIG. 26 illustrates a BER/SNR chart showing the performance differencebetween the SM scheme and the GC scheme using the outer code accordingto an embodiment of the present invention.

Specifically, FIG. 26 shows BER/SNR performance of the SC scheme and theGC scheme according to the code rate of the outer code under theassumption that a QPSK modulation scheme is used and channels are in aRayleigh channel environment. In charts described below, the term“Rayleigh channel environment” refers to an environment in whichchannels have no correlation between paths when MIMO transmission andreception is performed.

From FIG. 26, it can be seen that the SM scheme exhibits higherperformance than the GC scheme at a low code rate (¼, ⅓, ⅖, ½) having alarge minimum Hamming distance. However, it can also be seen that the GCscheme exhibits higher performance than the SM scheme at a high coderate (⅔, ¾, ⅘, ⅚) having a small minimum Hamming distance since thetransmit diversity gain of the GC scheme is high compared to performanceimprovement due to coding.

FIG. 27 illustrates BER/SNR charts showing the performance differencebetween the SM scheme and the GC scheme according to the code rate ofthe outer code and the modulation scheme according to an embodiment ofthe present invention.

The chart 203010 of FIG. 27 shows the case in which an outer code havinga code rate of ½ and a QPSK modulation scheme are used, the chart 203020shows the case in which an outer code having a code rate of ¾ and a QPSKmodulation scheme are used, and the chart 203030 shows the case in whichan outer code having a code rate of ⅚ and a 64-QAM modulation scheme areused.

From comparison of the charts 203010 to 203030, it can be seen that theSM scheme exhibits higher performance than the GC scheme when a low coderate (½) is used as shown in the chart 203010 and when a large QAM size(64-QAM) is applied as shown in the chart 203030.

Accordingly, the present invention suggests that a more efficient MIMObroadcast system be designed using a robust outer code while using an SMscheme having low complexity. However, the SM scheme may have a problemassociated with reception signal reconstruction (or restoration)depending on the degree of correlation between a plurality of MIMOtransmission and reception channels.

FIG. 28 illustrates a data transmission and reception method accordingto MIMO transmission of the SM scheme in a channel environment accordingto an embodiment of the present invention.

The MIMO transmission system may transmit input signal 1 (S1) and inputsignal 2 (S2) respectively through transmit antenna 1 and transmitantenna 2 according to the SM scheme. FIG. 28 illustrates an embodimentin which the transmitting side transmits a symbol modulated according to4-QAM.

The reception antenna 1 receives a signal through two paths. In thechannel environment of FIG. 28, the received signal of the receiveantenna 1 is S1*h₁₁+S2h₂₁ and the received signal of the receive antenna2 is S1*h₁₂+S2h₂₂. The receiving side may acquire S1 and S2 throughchannel estimation to reconstruct data.

This is a scenario in which the transmission and reception paths areindependent of each other. In the following, such an environment isreferred to as being uncorrelated. On the other hand, channels of thetransmission and reception paths may have a very high correlation witheach other as in a Line Of Sight (LOS) environment, which is referred toas being fully correlated in the following description.

In the case in which channels are fully correlated in MIMO, each channelmay be represented by a 2×2 matrix whose elements are all 1 (i.e.,h₁₁=h₁₂=h₂₁=h₂₂=1) as shown in FIG. 28. Here, the receive antenna 1 andthe receive antenna 2 receive the same reception signal (S1+S2). Thatis, if signals transmitted through 2 transmit antennas pass through thesame channel and are received by 2 receive antennas, a reception signalreceived by the receiver, i.e., data added (or combined) through thechannel, cannot express both symbols S1 and S2. As shown in FIG. 27, inthe fully correlated channel environment, the receiver cannot receive a16-QAM symbol, into which the signal S1 represented by a 4-QAM symboland the signal S2 represented by a 4-QAM symbol are combined and thereceiver cannot separate and reconstruct the signals S1 and S2 since thereceiver receives a combined signal S1+S2 represented by 9 symbols asshown on the right side of FIG. 28.

In this case, the receiver cannot reconstruct a signal receivedaccording to MIMO using the SM scheme even when the receiver is in avery high SNR environment. In the case of a communication system,communication is generally performed in two ways and therefore such achannel environment may be signaled to the transmitter through afeedback channel established between the transmitter and the receiver toallow the transmitter to change the transmission method. However, in thecase of a broadcast system, it may be difficult to perform bidirectionalcommunication through a feedback channel and one transmitter covers alarge number of receivers and a large range and therefore it may bedifficult to deal with various channel environment changes. Accordingly,if the SM scheme is used in such a fully correlated channel environment,the receiver cannot receive services and it is difficult to deal withsuch an environment, increasing costs, unless the coverage of thebroadcast network is reduced.

The following is a description of a method for dealing with the case inwhich the correlation between MIMO channels is 1, i.e., the case inwhich channels are in a fully correlated channel environment.

The present invention suggests that a MIMO system be designed such thatsignals received through MIMO channels satisfy the following conditionsso as to deal with the case in which the MIMO channels are fullycorrelated.

1) A received signal should be able to represent both original signalsS1 and S2. That is, coordinates of a constellation received by thereceiver should be able to uniquely represent sequences of S1 and S2.

2) A minimum Euclidean distance of a received signal should be increasedso as to reduce symbol error rate.

3) Hamming distance characteristics of a received signal should be goodso as to reduce bit error rate.

First, the present invention suggests a MIMO encoding method that uses aMIMO encoding matrix including an encoding factor “a” as expressed inthe following Expression 2 so as to satisfy such requirements.

$\begin{matrix}\begin{bmatrix}1 & a \\a & {- 1}\end{bmatrix} & \left\lbrack {{Expression}\mspace{14mu} 2} \right\rbrack\end{matrix}$

When a MIMO encoder encodes input signals S1 and S2 using a MIMOencoding matrix as shown in Expression 2, reception signal 1 (Rx1) andreception signal 2 (Rx2) received by antenna 1 and antenna 2 arecalculated as expressed in the following Expression 3. The receptionsignal 1 (Rx1) and reception signal 2 (Rx2) are calculated as expressedin the last line of Expression 3, especially, when MIMO channels arefully correlated.

Rx ₁ h ₁₁(S1+aS2)+h ₂₁(aS1−S2)

Rx ₂ =h ₁₂(S1+aS2)+h ₂₂(aS1−S2),

if h ₁₁ =h ₂₁ =h ₁₂ =h ₂₂ =h,

R=Rx ₁ =Rx ₂ =h{(a+1)S1+(a−1)S2}  [Expression 3]

First, when MIMO channels are uncorrelated, the reception signal 1 (Rx1)is calculated as Rx1=h₁₁(S1+a*S2)+h₂₁(a*S1−S1) and the reception signal2 (Rx2) is calculated as Rx2=h₁₂(S1+a*S2)+h₂₂(a*S1−S2). Thus, since thesignals S1 and S2 have the same power, it is possible to use gain of theMIMO system together with the SM scheme. When MIMO channels are fullycorrelated, the reception signals (R=Rx1=Rx2) expressed asR=h{(a+1)S1+(a−1)S2} are acquired and therefore it is possible toseparate and acquire the signals S1 and S2 and the signals S1 and S2 aredesigned such that both have different power and therefore it ispossible to secure robustness accordingly.

That is, the MIMO encoder may encode input signals S1 and S2 such thatthe input signals S1 and S2 have different powers according to theencoding factor “a” and are also received with different distributionseven in fully correlated channels. For example, input signals S1 and S2may be encoded such that both have different powers and the encodedsignals may then be transmitted using constellations which havedifferent Euclidean distances through normalization to allow thereceiver to separate and reconstruct the input signals even when thesignals have passed through fully correlated channels.

The MIMO encoding matrix described above may be represented asExpression 4 taking into consideration a normalization factor.

$\begin{matrix}{{\frac{1}{\sqrt{1 + a^{2}}}\begin{pmatrix}1 & a \\a & {- 1}\end{pmatrix}} = {\begin{pmatrix}\frac{1}{\sqrt{1 + a^{2}}} & \frac{a}{\sqrt{1 + a^{2}}} \\\frac{a}{\sqrt{1 + a^{2}}} & \frac{- 1}{\sqrt{1 + a^{2}}}\end{pmatrix} = \begin{bmatrix}{\cos \; \theta} & {\sin \; \theta} \\{\sin \; \theta} & {{- \cos}\; \theta}\end{bmatrix}}} & \left\lbrack {{Expression}\mspace{14mu} 4} \right\rbrack\end{matrix}$

MIMO encoding of the MIMO encoder that uses the MIMO encoding matrix (orrotation matrix) shown in Expression 4 may be considered as rotating theinput signals by an arbitrary angle of θ that can be represented by theencoding factor a, separating the cosine and sine components (or realand imaginary components) of the rotated signals, assigning positive andnegative (+/−) signs to the separated components, and transmitting theseparated components through different antennas. For example, the MIMOencoder may encode the input signals S1 and S2 such that the cosinecomponent of the input signal S1 and the sine component of the inputsignal S2 are transmitted through one transmit antenna and the sinecomponent of the input signal S and the cosine component of the inputsignal S2 to which a negative sign is attached are transmitted throughanother transmit antenna. The angle, by which the input signals arerotated, changes according to change of the value of the encoding factor“a” and the power distributions of the input signals S1 and S2 becomedifferent according to the value of the factor and the angle. Since thepower distribution difference can be represented by a distance betweensymbol coordinates in the constellations, the encoded input signals canbe represented by different constellations even when the input signalsare received by the receiving side via fully correlated channels suchthat it is possible to identify and separate the signals, therebyenabling reconstruction of the original input signals.

Specifically, the Euclidian distances of transmission signals change asthe power distributions change, the transmission signals received by thereceiving side can be represented by identifiable constellations havingdifferent Euclidian distances such that it is possible to reconstructthe signals even when the signals have passed through a fully correlatedchannel. That is, the MIMO encoder can encode the input signal S1 andthe input signal S2 into signals having different Euclidian distancesaccording to the value “a” and the receiving side can receive andreconstruct the encoded and transmitted signals using identifiableconstellations.

The MIMO encoder may perform encoding on input signals using the MIMOencoding matrix described above while additionally adjusting theencoding factor a. That is, it is possible to adjust and optimize theencoding factor “a” taking into consideration additional datareconstruction performance of the MIMO transmission and receptionsystem.

1. First Embodiment: MIMO Encoding Method that Optimizes the EncodingFactor “a” Taking into Consideration Euclidian Distances (in a FullyCorrelated MIMO Channel Environment)

It is possible to calculate the encoding factor value “a” taking intoconsideration the Euclidean distance while using the MIMO encodingmatrix. In a MIMO system having two transmit antennas and two receiveantennas, when transmission signal St1 is an M-QAM symbol andtransmission signal St2 is an N-QAM symbol, a signal St1+St2 that isreceived by the receiving side via a fully correlated MIMO channel is an(M*N)-QAM signal.

FIG. 29 illustrates input signals and transmission and reception signalswhen a MIMO encoding method has been performed according to anembodiment of the present invention.

In the embodiment of FIG. 29, an input signal S1 has a constellation205010 as a 4-QAM symbol and an input signal S2 has a constellation205020 as a 4-QAM symbol. When the input signal S1 and the input signalS2 are MIMO-encoded using the MIMO encoding matrix, the encodedtransmission signals St1 and St2 transmitted through antenna 1 (Tx1) andantenna 2 (Tx2) are 16-QAM symbols and have a constellation 205030 and aconstellation 205040 as shown in FIG. 29.

The first embodiment of the present invention suggests a method foroptimizing the value “a” such that symbols have the same Euclidiandistance in a constellation 205050 of a symbol of a reception signalthat has passed through a fully correlated channel as shown in FIG. 28.That is, in the case in which input signals are encoded using the MIMOmatrix, it is possible to calculate or set the value of the encodingfactor “a” such that reception symbols have the same Euclidean distancesin a constellation of a reception signal that has passed through a fullycorrelated channel and to encode the input signals using the calculatedor set value “a” of the encoding factor. Such a value “a” may berepresented by Expression 5 for each modulation scheme combination.

$\begin{matrix}{a = \left\{ \begin{matrix}{3,} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{\left( {4 + \sqrt{5}} \right)/\left( {4 - \sqrt{5}} \right)},} & {{{for}\mspace{14mu} {QPSK}} + {16\; Q\; A\; M}} \\{0.6,} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.} & \left\lbrack {{Expression}\mspace{14mu} 5} \right\rbrack\end{matrix}$

In the embodiment of FIG. 29, the constellation 205050 of the receptionsymbols corresponds to a constellation in which the value “a” has beenset to 3 and input signals have been MIMO-encoded through a combinationof 4-QAM and 4-QAM (i.e., QPSK+QPSK). That is, the distribution andconstellation of the transmission and reception symbols change accordingto modulation schemes of the reception signals and a combination of themodulation schemes and the Euclidean distance changes according to thedistribution and constellation of the symbols and therefore the value“a” for optimizing the Euclidean distance may also change accordingly.Expression 5 also shows an encoding factor value “a” for optimizing theEuclidean distance calculated when transmission and reception signalsare a combination of 4-QAM and 16-QAM (i.e., QPSK+16-QAM) and anencoding factor value “a” calculated when transmission and receptionsignals are a combination of 16-QAM and 16-QAM (i.e., 16-QAM+16-QAM).

FIG. 30 illustrates a BER/SNR chart showing the performance of the MIMOencoding method according to the first embodiment of the presentinvention.

Specifically, FIG. 30 shows a simulated performance difference betweenthe Golden Code (GC) scheme, the SM scheme, and the MIMO encoding method(SM OPT1) according to the first embodiment when transmission andreception signals are of 16-QAM in a fully correlated channel and 2transmit antennas and 2 receive antennas are provided. The followingcharts also show simulation results when an AWGN channel environmenthaving the same channel according to the MIMO transmission and receptionpaths is a fully correlated channel environment as in FIG. 30.

It can be seen from FIG. 3—that the MIMO encoding method according tothe first embodiment exhibits significantly better performance than theGC scheme or the SM scheme. In the chart of FIG. 30, an arrow shows anSNR gain of the first embodiment of the present invention. Specifically,it can be seen from FIG. 30 that the SNR gain increases as the code rateof the outer code increases. Especially, it can be seen that, in thecase of the SM scheme, decoding cannot be performed in a fullycorrelated MIMO channel environment and services cannot be receivedregardless of the SNR at a code rate higher than ⅖.

FIG. 31 illustrates a capacity/SNR chart showing the performance of theMIMO encoding method according to the first embodiment of the presentinvention in an uncorrelated channel.

In FIG. 31, a capacity satisfying a specific error rate on thehorizontal axis representing the SNR is shown according to each MIMOscheme. In the chart of FIG. 31, OSFBC denotes the Alamouti scheme. Fromthe chart, it can be seen that the MIMO encoding method of the firstembodiment of the present invention exhibits the same performance as theSM scheme while exhibiting the best performance among other schemes.

FIG. 32 illustrates a capacity/SNR chart showing the performance of theMIMO encoding method according to the first embodiment of the presentinvention in a fully correlated channel.

It can be seen from FIG. 32 that, in a fully correlated MIMO channel,the MIMO encoding method according to the first embodiment exhibitssignificantly better SNR performance than the SM scheme and the GCscheme and also exhibits better performance than the SISO scheme.

Accordingly, it can been from the charts of FIGS. 31 and 32, the MIMOencoding method according to the first embodiment of the presentinvention can achieve higher performance than the GC scheme while usinga system having lower complexity than the GC scheme and can also achievesignificantly better performance than the SM scheme having similarcomplexity in a fully correlated channel environment.

In another embodiment of the present invention, a GC subset may be usedas a MIMO encoding matrix when MIMO encoding is performed. In this case,the MIMO encoding matrix is represented by Expression 6.

$\begin{matrix}{{\begin{bmatrix}\alpha & {\alpha \; \theta} \\{i\; \overset{\_}{\alpha}} & {i\; \overset{\_}{\alpha}\; \overset{\_}{\theta}}\end{bmatrix}\begin{bmatrix}{S\; 1} \\{S\; 2}\end{bmatrix}},{\alpha = {1 + {\left( {1 - \theta} \right)i}}},{\overset{\_}{\alpha} = {1 + {\left( {1 - \overset{\_}{\theta}} \right)i}}},{\theta = \frac{1 + \sqrt{5}}{2}},{\overset{\_}{\theta} = \frac{1 - \sqrt{5}}{2}}} & \left\lbrack {{Expression}\mspace{14mu} 6} \right\rbrack\end{matrix}$

Using the encoding matrix of Expression 6 exhibits better performancethan the first embodiment of the present invention.

FIG. 33 illustrates a constellation when a GC subset is used as a MIMOencoding matrix and a constellation when the first embodiment isapplied.

The constellation of FIG. 33 is a constellation in the case in which a16-QAM type input signal S1 and 16-QAM type input signal S2 areMIMO-encoded using a MIMO encoding matrix and signals transmittedthrough 2 transmit antennas are received by a receiver through a fullycorrelated channel. The left part of FIG. 33 shows a receptionconstellation when a GC subset is used and the right part shows areception constellation when the first embodiment is used.

FIG. 34 illustrates a capacity/SNR chart showing a performancecomparison between when a GC subset is used as a MIMO encoding matrixand when the first embodiment is used.

As can be seen from the chart, SNR performance is higher when the GCsubset is used while the minimum Euclidean distance of a constellationof a reception signal when the first embodiment (SM OPT1) is applied isgreater than when the GC subset is used. Thus, the performance of thefirst embodiment differs due to a factor other than the Euclidiandistance, the reason of which is described below.

FIG. 35 illustrates a relationship between Euclidean distance andHamming distance in a constellation when a GC subset is used as a MIMOencoding matrix and in a constellation when the first embodiment isused.

The reason why the SNR performance of the first embodiment is lower thanthat when the GC subset is used although the minimum Euclidean distanceof the first embodiment is greater than when the GC subset is used isassociated with the relationship between the Euclidian distance and theHamming distance.

Hamming distance distributions when the first embodiment is applied andwhen the GC subset is used are similar and have no gray mapping.However, it can be seen from FIG. 35 that the Euclidian distance of agreen line pair or a black line pair having a greater Hamming distancewhen the GC subset is used is greater than that when the firstembodiment is applied. That is, although internal Euclidian distances of4 by 4 16-QAM constellations which are distributed over 16 areas in thetotal constellation are similar in both cases, the Euclidian distancebetween the 4 by 4 16-QAM constellations when the GC subset is used isgreater, thereby compensating for the Hamming distance performancedifference.

Due to such characteristics, the case of using the GC subset exhibitshigher BER performance than the case of the first embodiment althoughthe minimum Euclidean distance when the GC subset is used is smallerthan when the first embodiment is applied. Accordingly, in thefollowing, the present invention suggests a MIMO encoding method havinghigher SNR performance or BER performance.

2. Second Embodiment: MIMO Encoding Method Taking into ConsiderationGray Mapping in Addition to Euclidian Distance

The second embodiment suggests a MIMO encoding method in which anencoding factor value “a” is set so as to optimize the Euclideandistance, similar to the first embodiment, and MIMO encoding isperformed such that a reception signal that has passed through a fullycorrelated channel has a gray mapping (or gray mapping form).

In the MIMO encoding method of the second embodiment, at the receivingside, the signs of real and imaginary parts of the input signal S2 amongthe input signals S1 and S2 may be changed according to a value of theinput signal S1 such that each signal becomes a gray mapping signal.Data values included in the input signal S2 may be changed using amethod represented by the following Expression 7.

  [Expression 7] S1 = b₀b₁...b_(N−1), N = log₂ M, M = QAM size of S1real(S1) = b₀b₂..b_(N−2) imag(S1) = b₁b₃..b_(N−1)  for i = 1...N − 1  si = sq = 1   if i = index of real(S1) and b₁ = 1    si = −si   if i =index of imag(S1) and b₁ = 1    sq = −sq  end for  S2 = si · real(S2) +i · sq · imag(S2)

FIG. 36 illustrates input signals and transmission and reception signalswhen a MIMO encoding method has been performed according to the secondembodiment of the present invention.

If bit values assigned to the real and imaginary parts of the inputsignal S1 212010 among the input signals S1 and S2 212010 and 212020 areXORed as in Expression 7 and the signs of the real and imaginary partsare determined according to the XORed value and transmission signal 1202030 and transmission signal 2 212040 are transmitted respectivelythrough antenna 1 and antenna 2, then reception symbols of a receptionsignal 212050, which is received by the receiver via a fully correlatedchannel, have a gray mapping form such that the Hamming distance betweenadjacent symbols in the constellation does not exceed 2 as shown in FIG.36.

Since an (M*N)-QAM signal received by the receiver has a minimumEuclidean distance and a gray mapping form, the second embodiment mayachieve the same performance as the SIMO scheme even in a fullycorrelated MIMO channel environment. However, when signals S1 and S2 areacquired by decoding the reception signal at the ML decoder, complexitymay be increased since the value of S2 depends on the value of S1 andperformance may be degraded due to the correlation between input signalsin an uncorrelated MIMO channel.

3. Third Embodiment: MIMO Encoding Method that Sets MIMO Encoding FactorTaking into Consideration Hamming Distance in Addition to EuclidianDistance

The third embodiment suggests a method in which MIMO encoding isperformed by setting an encoding factor value “a” so as to optimize theEuclidian distance taking into consideration the Hamming distance of areception signal rather than allowing the entire constellation of thereception signal to have a Euclidian distance as in the firstembodiment.

FIG. 37 illustrates a MIMO encoding method according to the thirdembodiment of the present invention.

FIG. 37 illustrates a relationship between the value of an encodingfactor “a” of a MIMO encoding matrix and a Hamming distance in aconstellation of a reception signal received through a fully correlatedMIMO channel. In the third embodiment, a Hamming distance of intervalD_E1 is smaller than a Hamming distance of interval D_E2 in theconstellation of the reception signal and therefore the Euclidiandistance is adjusted so as to compensate for the Hamming distancedifference by maintaining the power difference between the interval D_E1and the interval D_E2 such that the power of the interval D_E1 is twicethe power of the interval D_E2. That is, the Euclidian distance isadjusted so as to compensate for the reconstruction performancedifference due to the Hamming distance difference using the powerdifference.

In the example of FIG. 37, the Hamming distance of the interval D_E2 istwice higher than that of the interval D_E1. That is, the Euclidiandistance between adjacent symbols in an interval, whose Hamming distanceis twice greater than another interval since the number of bits thereofis twice greater than the other interval, can be increased so as toincrease power of the interval, thereby compensating for performancedegradation due to the Hamming distance difference when a receptionsignal is reconstructed. First, a relative Euclidian distance of areception signal into which 2 transmission signals St1 and St2 receivedby the receiver are combined as shown in FIG. 36 is determined. It canbe seen from the above Expression 1 that the minimum Euclidean distanceof a 16-QAM symbol whose power is reduced is 2(a−1) and the minimumEuclidean distance of a 16-QAM symbol whose power is increased is 2(a+1)(since one reception signal is expressed as R=h{(a+1)S1+(a−1)S2}). Itcan be seen from FIG. 36 that d_E1 is equal to the Euclidian distance of16-QAM symbols whose power has been reduced. It can also be seen fromFIG. 20 that D_E2 is twice a distance obtained by subtracting 3/2 of theEuclidean distance of 16-QAM symbols whose power has been reduced from ½of the Euclidean distance of 16-QAM symbols whose power has beenincreased. This may be represented by Expression 8.

$\begin{matrix}{{2D_{H_{1}}} = D_{H_{2}}} & \left\lbrack {{Expression}\mspace{14mu} 8} \right\rbrack \\{{\sqrt{2}D_{E_{1}}} = D_{E_{2}}} & \; \\{{2\sqrt{2}\left( {a - 1} \right)} = {2\left( {\left( {a + 1} \right) - {3\left( {a - 1} \right)}} \right)}} & \; \\{a = \frac{\sqrt{2} + 4}{\sqrt{2} + 2}} & \;\end{matrix}$

That is, the MIMO encoder performs MIMO encoding on input signals bydistributing different powers to the input signals using the MIMO matrixsuch that the signals have different Euclidian distances. In this case,the MIMO encoder may perform MIMO encoding by calculating and settingthe encoding factor value “a” such that input signals with distributedpower have Euclidian distances for compensating for a Hamming distancedifference according to the third embodiment.

FIG. 38 illustrates input signals and transmission and reception signalswhen a MIMO encoding method has been performed according to the thirdembodiment of the present invention.

In the example of FIG. 38, when an input signal S1 (214010) and an inputsignal S2 (214020) are MIMO-encoded according to the third embodiment,the encoded transmission signals have constellations (214030) and(214040). When the transmission signals are transmitted through a MIMOchannel, a reception signal received by the receiver has a constellation214050. It can be seen from the constellation of the reception signal214050 that the Euclidean distance has been adjusted according to theHamming distance.

In the example described above with reference to FIGS. 37 and 38, thevalue “a” is calculated when the input signal S1 is a 16-QAM signal andthe input signal S2 is also a 16-QAM signal. The value “a” of adifferent modulation scheme may be calculated as shown in Expression 8using the same principle.

                                [Expression  9]$a = \left\{ \begin{matrix}{{\sqrt{2} + 1},} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{\left( {\sqrt{2} + 3 + \sqrt{5}} \right)/\left( {\sqrt{2} + 3 - \sqrt{5}} \right)},} & {{{for}\mspace{14mu} {QPSK}} + {16\; Q\; A\; M}} \\{{\left( {\sqrt{2} + 4} \right)/\left( {\sqrt{2} + 2} \right)},} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.$

Here, it is assumed that, in the case of QPSK+16-QAM MIMO, the valuessuggested above are obtained when power of the input signals S1 and S2has been normalized to 1 after the input signals S1 and S2 areQAM-modulated through QPSK and 16-QAM, respectively, at the symbolmapper. When the power has not been normalized, the value “a” may bemodified accordingly.

In addition, in the case of QPSK+16-QAM, a value of 4.0 other than theabove-suggested values may be used as the value “a”. The reason for thisis that the combined signal can represent all input signals S1 and S2even when the SM scheme is applied in a fully correlated channelenvironment. In this case, a value of 4.0 or a value close to 4.0 may beused instead of the value calculated using Expression 9 in order tocompensate a high code rate of the outer code.

FIG. 39 illustrates capacity/SNR charts showing a performance comparisonof MIMO encoding methods according to the present invention.

It can be seen from the left chart that, in a fully correlated MIMOchannel environment, the MIMO encoding method (SM OPT2) of the secondembodiment has almost the same performance as the SIMO scheme. However,it can be seen from the right chart that, in an uncorrelated MIMOchannel environment, the MIMO encoding method (SM OPT2) of the secondembodiment suffers from performance degradation due to the relationshipbetween the MIMO-encoded and transmitted signals S1 and S1,specifically, since the signal S2 depends on the signal S1 as describedabove.

It can also be seen that the MIMO encoding method (SM OPT3) of the thirdembodiment experiences no performance loss in an uncorrelated MIMOchannel while exhibiting performance better than the first embodiment(SM OPT1) in a fully correlated MIMO channel (or channel environment).

FIG. 40 illustrates different capacity/SNR charts showing a performancecomparison of MIMO encoding methods according to the present invention.

It can be seen from the left chart that, in a fully correlated MIMOchannel environment, the MIMO encoding method (SM OPT3) of the thirdembodiment exhibits better performance than the first embodiment (SMOPT1) and the MIMO encoding method (SM OLDP Golden) that uses a subsetof Gold Code (GC) and it can also be seen from the right chart that theMIMO encoding method (SM OPT3) of the third embodiment experiences noperformance loss in an uncorrelated MIMO channel environment.

When the second embodiment and the third embodiment are compared withreference to the above descriptions and charts, it can be seen that thesecond embodiment exhibits the same performance as SIMO in a fullycorrelated MIMO channel environment and thus does not suffer anyperformance loss, thereby solving the problems of the MIMO scheme in afully correlated MIMO channel environment. However, in the secondembodiment, input signals S1 and S2 are not independent of each otherdue to MIMO encoding such that the signal S2 changes according to thesignal S1, thereby causing performance degradation in an uncorrelatedchannel as can be seen from FIGS. 39 and 40. Accordingly, iterative MLdetection may be used in order to solve the problem that reception anddecoding errors of the signal S1 are reflected in the signal S2, causingan additional decoding error of the signal S2.

In the iterative ML detection method, an outer code is included in aniterative loop and a detection error of the signal S1 is reduced using asoft posteriori probability value of the signal S1 output from an outerport as an a priori probability value of the ML detector, therebyreducing the probability of application of the detection error of thesignal S1 for detection of the signal S2. This method allows the MIMOencoding method of the second embodiment to exhibit performance of theSIMO system in a fully correlated MIMO channel environment and exhibitperformance of the SIMO system in an uncorrelated MIMO channelenvironment.

In the MIMO encoding method of the third embodiment, a reception signalreceived through a fully correlated MIMO channel is designed taking intoconsideration both the Hamming distance and the Euclidian distance.Accordingly, the MIMO encoding method of the third embodiment not onlyhas better performance in a fully correlated MIMO channel but also hasno performance loss compared to the SM scheme in an uncorrelated MIMOchannel such that it is possible to use both MIMO transmission andreception gains. In this case, there is an advantage in implementationof the receiver since the receiver has complexity similar to the SMscheme.

FIG. 41 illustrates capacity/SNR charts showing a performance comparisonof combinations of modulation schemes in the MIMO encoding methodaccording to the third embodiment of the present invention.

Specifically, FIG. 41 shows a performance comparison of a QPSK+QPSK MIMOtransmission scheme and a 16-QAM+16-QAM MIMO transmission scheme of thethird embodiment with SIMO schemes of 16-QAM, 64-QAM, and 256-QAM andSISO schemes of 16-QAM, 64-QAM, and 256-QAM.

From the left chart, it can be seen that, in an uncorrelated channelenvironment, the third embodiment achieves the MIMO transmission andreception gain and exhibits significantly better performance than theSIMO or SISO scheme. From the right chart, it can also be seen that, ina fully correlated channel environment, the third embodiment exhibitsbetter performance than the SISO scheme but there is a performancedifference between the QPSK+QPSK MIMO transmission scheme and the16-QAM+16-QAM MIMO transmission scheme as shown. A QPSK+16-QAMtransmission scheme may be used to compensate for the performancedifference. In the QPSK+16-QAM transmission scheme, data of one of theinput signals S1 and S2 used for MIMO encoding/decoding is a QPSK symboland data of the other is a 16-QAM symbol. In this case, the amount ofdata that is transmitted at once is similar to 64-QAM of the SISOscheme.

FIG. 42 illustrates capacity/SNR charts showing a performance comparisonof different channel correlation levels when a QPSK+QPSK MIMOtransmission scheme is used in the MIMO encoding method according to thethird embodiment of the present invention.

The charts of FIG. 42 show measured performance of different MIMOchannel correlation levels. A range from (cor 0.0) corresponding to acorrelation level of 0 to (cor 1.0) corresponding to a correlation levelof 1 is divided into 0.0, 0.3, 0.5, 0.7, 0.9, and 1.0 and performance ofeach correlation level is shown in a corresponding chart.

From the charts of FIG. 42, it can be seen that, when the encodingmethod of the third embodiment uses a QPSK+QPSK MIMO transmissionscheme, performance increases as the correlation between channelsincreases. It can also be seen that performance is degraded to theextent that decoding is not possible in the case of a fully correlatedMIMO channel (cor 1.0) when the SM scheme is used.

When the GC scheme is used, the encoding method may exhibit performancewhich increases as the code rate increases. However, the increasedperformance may still be small and the method exhibits performance lowerthan the embodiments of the present invention at a low code rate. Fromthe chart of FIG. 42, it can be seen that the performance of the GCscheme is seriously degraded in a fully correlated MIMO channelenvironment.

FIG. 43 illustrates capacity/SNR charts showing a performance comparisonof different channel correlation levels when a QPSK+16-QAM MIMOtransmission scheme is used in the MIMO encoding method according to thethird embodiment of the present invention.

The charts of FIG. 43 show measured performance of different MIMOchannel correlation levels. A range from (cor 0.0) corresponding to acorrelation level of 0 to (cor 1.0) corresponding to a correlation levelof 1 is divided into 0.0, 0.3, 0.5, 0.7, 0.9, and 1.0 and performance ofeach correlation level is shown in a corresponding chart.

From the charts of FIG. 43, it can be seen that, when the encodingmethod of the third embodiment uses a QPSK+16-QAM MIMO transmissionscheme, performance increases as the correlation between channelsincreases. It can also be seen that performance is significantlydegraded in a fully correlated MIMO channel (cor 1.0) when the SM schemeor the GC scheme is used.

FIG. 44 illustrates capacity/SNR charts showing a performance comparisonof different channel correlation levels when a 16-QAM+16-QAM MIMOtransmission scheme is used in the MIMO encoding method according to thethird embodiment of the present invention.

The charts of FIG. 44 show measured performance of different MIMOchannel correlation levels. A range from (cor 0.0) corresponding to acorrelation level of 0 to (cor 1.0) corresponding to a correlation levelof 1 is divided into 0.0, 0.3, 0.5, 0.7, 0.9, and 1.0 and performance ofeach correlation level is shown in a corresponding chart.

From the charts of FIG. 44, it can be seen that, when the encodingmethod of the third embodiment uses a 16-QAM+16-QAM MIMO transmissionscheme, performance increases as the correlation between channelsincreases. It can also be seen that performance is significantlydegraded in a fully correlated MIMO channel (cor 1.0) when the SM schemeor the GC scheme is used. Especially, it can be seen that, when the SMscheme is used, decoding is not possible at all code rates in a fullycorrelated MIMO channel environment.

Power imbalance may occur between signals that are transmitted throughrespective communication paths when MIMO transmission and reception isperformed. That is, signals transmitted through a plurality oftransmission antennas may be received by a receiver with differentpowers. In the worst case, only a signal transmitted by one transmissionantenna may be received by the receiver. The following is a descriptionof a MIMO encoding method which can minimize performance degradation insuch a power imbalance situation.

4. Fourth Embodiment: MIMO Encoding Method that Optimizes the EncodingFactor “a” Taking into Consideration Euclidian Distances of TransmissionSignals

It is possible to calculate the encoding factor value “a” taking intoconsideration the Euclidean distance while using the MIMO encodingmatrix. In a MIMO system having two transmit antennas and two receiveantennas, when transmission signal St1 is an M-QAM symbol andtransmission signal St2 is an N-QAM symbol, a signal St1+St2 that isreceived by the receiving side via a fully correlated MIMO channel is an(M*N)-QAM signal.

FIG. 45 illustrates input signals and transmission signals on which aMIMO encoding method according to a fourth embodiment of the presentinvention has been performed.

In the embodiment of FIG. 45, the input signal S1 has a constellation221010 as a 16-QAM symbol and the input signal S2 has a constellation221020 as a 16-QAM symbol. If the input signal S1 and the input signalS2 are MIMO-encoded using the MIMO encoding matrix, the encodedtransmission signals St1 and St2 transmitted through antenna 1 (Tx1) andantenna 2 (Tx2) are 256-QAM symbols and the constellations of theencoded transmission signals have no minimum Euclidean distance.Although the value “a” may be determined by optimizing the Euclidiandistance with reference to a reception signal, this may degrade decodingperformance in a power imbalance situation. Accordingly, if theEuclidian distance is optimized with reference to transmission signalsfrom the transmitting side, it is possible to minimize performancedegradation that occurs in a power imbalance situation at the receivingside. Optimization of Euclidean distance means arrangement of symbols atan equal interval in a signal constellation and maximization of aminimum Euclidean distance in the constellation. The value “a” foroptimizing the Euclidian distance with reference to transmission signalsmay be represented by Expression 10.

$\begin{matrix}{a = \left\{ \begin{matrix}{2,} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{4,} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.} & \left\lbrack {{Expression}\mspace{14mu} 10} \right\rbrack\end{matrix}$

When the value “a” determined according to Expression 10 is used, i.e.,when the value “a” is set to 4 and MIMO encoding is performed using thesame in the example of expressions 1 or 3, the transmission signals St1and St2 have a constellation 221030 and a constellation 221040,respectively. From the constellations 221030 and 221040 of thetransmission signals St1 and St2, it can be seen that the Euclidiandistance is distributed uniformly. Accordingly, it can be understoodthat, when the MIMO encoding method of the first embodiment is used, itis possible to minimize reception performance degradation since theEuclidian distance of the transmission signals received in a powerimbalance situation has been optimized.

However, a different value needs to be used when symbol types of inputsignals are different as in the QPSK+16-QAM transmission method. Thereason for this is that, when modulation schemes of input signals usedfor MIMO transmission are different, a trade-off problem occurs suchthat, if the optimized value “a” is used for one transmission antenna,then a signal having relatively low performance is transmitted throughanother transmission antenna.

5. Fifth Embodiment: MIMO Encoding Method Taking into Consideration GrayMapping in Addition to Euclidian Distance

The fifth embodiment suggests a MIMO encoding method in which anencoding factor value “a” is set so as to optimize the Euclideandistance, similar to the first embodiment, and MIMO encoding isperformed such that a reception signal that has passed through a fullycorrelated channel has a gray mapping (or gray mapping form).

In the MIMO encoding method of the second embodiment, at the receivingside, the signs of real and imaginary parts of the input signal S2 amongthe input signals S1 and S2 may be changed according to a value of theinput signal S1 such that each signal becomes a gray mapping signal.Data values included in the input signal S2 may be changed using amethod represented by the following Expression 7.

That is, the MIMO encoder may perform MIMO encoding after changing signsof the input signal S2 according to the value of the input signal S1while using the same MIMO encoding factor as used in the firstembodiment. In other words, the sign of the input signal S2 may bedetermined according to the sign of the input signal S1, and then theMIMO encoding matrix may be applied to the first and second inputsignals S1 and S2 to output the first and second transmission signals,as described above.

Since an (M*N)-QAM signal received by the receiver has a minimumEuclidean distance and a gray mapping form, the second embodiment mayachieve the same performance as the SIMO scheme even in a fullycorrelated MIMO channel environment. However, when signals S1 and S2 areacquired by decoding the reception signal at the ML decoder, complexitymay be increased since the value of S2 depends on the value of S1 andperformance may be degraded due to the correlation between input signalsin an uncorrelated MIMO channel.

6. Sixth Embodiment: MIMO Encoding Method that Sets MIMO Encoding FactorTaking into Consideration Hamming Distance in Addition to EuclidianDistance

The sixth embodiment suggests a method in which MIMO encoding isperformed by setting an encoding factor value “a” so as to optimize theEuclidian distance taking into consideration the Hamming distance of areception signal rather than allowing the entire constellation of thereception signal to have a Euclidian distance as in the firstembodiment. Specifically, it is possible to design an encoding matrixsuch that the square of the Euclidian distance between adjacentconstellation points when the Hamming distance between the adjacentconstellation points is 2 is twice the square of the Euclidian distancebetween the points when the Hamming distance is 1. That is, theEuclidian distance is adjusted such that a reconstruction performancedifference due to a Hamming distance difference can be compensated forusing a power difference.

Assuming that FIG. 37 shows a constellation of a transmit signal, aHamming distance of the interval D_E1 is a half of a Hamming distance ofthe interval D_E2 in the constellation of the transmission signal, andtherefore the Euclidian distance is adjusted so as to compensate for thehamming distance difference by maintaining the power difference betweenthe interval D_E1 and the interval D_E2 such that the power of theinterval D_E1 is twice the power of the interval D_E2. That is, theHamming distance of the interval D_E2 is twice higher than that of theinterval D_E1. That is, the Euclidian distance between adjacent symbolsin an interval, whose Hamming distance is twice greater than anotherinterval since the number of bits thereof is twice greater than theother interval, can be increased so as to increase power of theinterval, thereby compensating for performance degradation due to theHamming distance difference when a reception signal is reconstructed.

The value “a” may be obtained using the above conditions, which may berepresented by Expression 7.

$\begin{matrix}{a = \left\{ \begin{matrix}{{\sqrt{2} + 1},} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{\sqrt{2} + 3},} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.} & \left\lbrack {{Expression}\mspace{14mu} 11} \right\rbrack\end{matrix}$

FIG. 46 illustrates input signals and transmission signals on which aMIMO encoding method according to a sixth embodiment of the presentinvention has been performed.

In the embodiment of FIG. 46, an input signal S1 corresponds to 16-QAMsymbols and has a constellation 207010, whereas an input signal S2corresponds to 16-QAM symbols and has a constellation 207020. When theinput signal S1 and the input signal S2 are encoded using a MIMOencoding matrix including an encoding parameter set according to thethird embodiment, encoded transmission signals St1 and St2, which aretransmitted through antenna 1 (Tx1) and antenna 2 (Tx2), become 256-QAMsymbols and respectively have constellations 207030 and 207040 havingEuclidean distances taking Hamming distance into account. Furthermore,the transmission signals have symbol distributions having the Euclideandistances taking the hamming distance into account, and thus performanceloss can be minimized even when a receiver receives and decodes only oneof the transmission signals in a power imbalance situation.

However, when the input signals have different symbol types as in aQPSK+16-QAM transmission method, it is necessary to use different values‘a’. This is because, if a value ‘a’ optimized for one of thetransmission antennas is used, a signal having lower quality may betransmitted through the other transmission antenna when differentmodulation schemes are employed for the input signals used for MIMOtransmission, that is, trade-off may be generated.

Among the above-mentioned MIMO encoding schemes, the precoding matrixcan be used to improve throughput or performance of the SM schemeaccording to one embodiment of the present invention. In the case ofusing the precoding matrix, input signals S1 and S2 associated with allcorrelation channels can be represented through Antenna 1 and Antenna 2,respectively, resulting in an increased diversity. Detailed precodingmatrices are represented by Equation 4 and Equations 12 to 14.

$\begin{matrix}{{\frac{1}{\sqrt{a^{2} + 1}}\begin{bmatrix}1 & {- a} \\a & 1\end{bmatrix}}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}} & \left\lbrack {{Equation}\mspace{14mu} 12} \right\rbrack \\{{\begin{bmatrix}{\cos \; \theta} & {{- \sin}\; \theta} \\{\sin \; \theta} & {\cos \; \theta}\end{bmatrix}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{\theta = {{atan}(a)}}} & \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack \\{{\begin{bmatrix}{\cos \; {\theta (n)}} & {{- \sin}\; {\theta (n)}} \\{\sin \; {\theta (n)}} & {\cos \; {\theta (n)}}\end{bmatrix}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{{\theta (n)} = {\frac{2\; \pi}{N}n}},{n = 0},\ldots \mspace{14mu},{N - 1}} & \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack\end{matrix}$

Equations 12 and 13 are different from Equation 4 in terms ofrepresentation, but the Equations 12 and 13 have the same mathematicalmeaning as that of Equation 4. In Equation 14, a rotation angle of thematrix is not fixed, is changed according to the value of n, such thatthe changed result can be used. In this case, n may be changed accordingto an OFDM carrier index or an OFDM symbol index. Here, after thereceiver is synchronized with the transmitter to obtain the value of n,the receiver may perform decoding. In addition, the embodiment based onEquation 14 does not follow MIMO encoding performance given according toa specific rotation angle of the matrix, and may follow averageperformance of several rotation angles.

In order to improve performance of the above-mentioned SM scheme, amatrix may be used in consideration of the precoding matrix and thephase rotation according to one embodiment of the present invention.

In this case, influence caused by channel variation according totransmitter characteristics can be minimized, and additional diversitycan be obtained. In addition, a phase rotation may also be applied onlyto a path of any one of several antennas. This example may be changedaccording to a designer's intention.

Equation 15 is a matrix obtained when phase rotation is applied to theprecoding matrix.

$\begin{matrix}{{{\begin{bmatrix}1 & 0 \\0 & e^{j\; {\varphi {(k)}}}\end{bmatrix}\begin{bmatrix}{Precoding} \\{Matrix}\end{bmatrix}}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{{\varphi (k)} = {\frac{2\; \pi}{N}k}},{k = 1},\ldots \mspace{14mu},{N - 1}} & \left\lbrack {{Equation}\mspace{14mu} 15} \right\rbrack\end{matrix}$

The precoding matrix located at the center part of Equation 15 may beset to any one of Equation 4 and Equations 4 to 14. In the case of usingthe matrix of Equation 15, signals transmitted from Antenna 1 andAntenna 2 may represent Signal S1 and Signal S2 as necessary. Therefore,the transmitter minimizes channel variation influence caused bytransmitter characteristics, and transmits signals using the minimizedchannel variation. The receiver may separate the signals S1 and S2 fromeach other using only signals transmitted from the transmitter, and mayrecover each of the signals S1 and S2. In more detail, the receiver maysynchronize and obtain the index K of the phase rotation used for thematrix, and may calculate LLR using the precoding matrix.

Equation 15 may also be represented by Equation 16 and Equation 17.

$\begin{matrix}{\mspace{79mu} {{{\begin{bmatrix}1 & 0 \\0 & e^{j\; {\varphi {(k)}}}\end{bmatrix}\begin{bmatrix}{\cos \; \theta} & {{- \sin}\; \theta} \\{\sin \; \theta} & {\cos \; \theta}\end{bmatrix}}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{\theta = \left\{ \begin{matrix}{{{atan}\left( {\sqrt{2} + 1} \right)},} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{{atan}\left( {\left( {\sqrt{2} + 4} \right)/\left( {\sqrt{2} + 2} \right)} \right)},} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.}}} & \left\lbrack {{Equation}\mspace{14mu} 16} \right\rbrack\end{matrix}$

Equation 16 may also be denoted by the following Equation 17.

$\begin{matrix}{{{{\frac{1}{\sqrt{a^{2} + 1}}\mspace{11mu}\begin{bmatrix}1 & 0 \\0 & e^{j\; {\varphi {(k)}}}\end{bmatrix}}\begin{bmatrix}1 & a \\a & {- 1}\end{bmatrix}}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{{\varphi (k)} = {\frac{2\; \pi}{N}k}},{k = 1},\ldots \mspace{14mu},{N - 1},\left\{ \begin{matrix}{{\sqrt{2} + 1},} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{\left( {\sqrt{2} + 4} \right)/\left( {\sqrt{2} + 2} \right)},} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 17} \right\rbrack\end{matrix}$

Equations 16 and 17 are different from Equation 4 in terms ofrepresentation, but the Equations 16 and 17 have the same mathematicalmeaning. Equation 16 indicates an optimum value 0 at which influence ofa channel variation is minimized when input signals respectivelycorrespond to “4-QAM and 4-QAM” (i.e., QPSK+QPSK) and “16-QAM and16-QAM”.

Equation 15 can also be represented by Equation 18.

$\begin{matrix}{\; {{{\begin{bmatrix}1 & 0 \\0 & e^{j\; {\varphi {(k)}}}\end{bmatrix}\begin{bmatrix}{\cos \; \theta} & {{- \sin}\; \theta} \\{\sin \; \theta} & {\cos \; \theta}\end{bmatrix}}\begin{bmatrix}s_{1} \\s_{2}\end{bmatrix}},{\theta = \left\{ {\begin{matrix}{{29\mspace{14mu} {degrees}},} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{{16.8\mspace{14mu} {degrees}},} & {{{for}\mspace{14mu} 16Q\; A\; M} + {16Q\; A\; M}}\end{matrix},{{\varphi (k)} = {\frac{2\; \pi}{N}k}},{k = 1},\ldots \mspace{14mu},{N - 1}} \right.}}} & \left\lbrack {{Equation}\mspace{14mu} 18} \right\rbrack\end{matrix}$

A signal that is MIMO-encoded according to the MIMO encoding matrixshown in Equation 18 has robustness against Rayleigh fading, and theinput signal S1 and the other input signal S2 can be separated from eachother even in all correlation channels.

FIG. 47 is a flowchart illustrating a method of transmitting a broadcastsignal according to an embodiment of the present invention.

A transmitter for transmitting a broadcast signal according to oneembodiment of the present invention generates a first input signal and asecond input signal for MIMO transmission in step S47100.

A first input signal and a second input signal are generated by theinput signal generator or the divider. The broadcast transmitter maydivide data to be transmitted into the first input signal and the secondinput signal according to a MIMO transmission path. The input signalgenerator or the divider may perform unique operations using differentdevice elements.

The transmitter for transmitting a broadcast signal performs MIMOencoding of the first input signal and the second input signal, andgenerates the first transmission signal and the second transmissionsignal in step S47200.

The MIMO encoding operation may be carried out by the MIMO encoder orthe MIMO processor as described above, and may use the MIMO encodingmatrix corresponding to the above-mentioned embodiments. That is, thebroadcast signal transmitter may perform MIMO encoding using either MIMOmatrices described in FIG. 25 to 46 or MIMO matrices described inEquations 12 to 18. In more detail, a signal obtained by addition of thefirst transmission signal and the second transmission signal may have anoptimized Euclidean distance, and may have an Euclidean distance that iscapable of having gray mapping or compensating for the Hamming distance,such that MIMO encoding may be carried out using the MIMO matrix atwhich the parameter ‘a’ is established. As described above, thebroadcast transmitter may adjust power of input signals using the MIMOmatrix, where the parameter ‘a’ may be set to different values accordingto modulation types of input signals.

MIMO encoding may be carried out using the above-mentioned precodingMIMO matrices shown in Equations 12 to 14, and MIMO encoding may becarried out using the phase-rotated MIMO matrices shown in Equations 15to 18. This example may be changed according to designer intention.Thereafter, the broadcast transmitter may perform OFDM modulation of thefirst transmission signal and the second transmission signal, and maytransmit the OFDM-modulated first and second transmission signals instep S473000.

The broadcast transmitter may transmit the OFDM-modulated firsttransmission signal and the OFDM-modulated second transmission signalthrough the first antenna and the second antenna, respectively. Thefirst transmission signal and the second transmission signal may havethe above-mentioned signal frame structures.

FIG. 48 is a flowchart illustrating a method of receiving a broadcastsignal according to an embodiment of the present invention.

Referring to FIG. 48, the broadcast receiver according to one embodimentof the present invention receives a first reception signal and a secondreception signal, and performs OFDM modulation of the first receptionsignal and the second reception signal in step S48000. The firstreception signal and the second reception signal are MIMO-encoded by thebroadcast transmitter according to one embodiment of the presentinvention, and the MIMO encoding operation may be carried out by theMIMO encoder or the MIMO processor as described above, and the MIMOencoding matrices corresponding to the above-mentioned embodiments maybe used. That is, the broadcast transmitter may perform MIMO encodingusing any one of the MIMO matrices described in Equations 12 to 18 andMIMO matrices described in Equations 25 to 46. In more detail, a signalobtained by addition of the first transmission signal and the secondtransmission signal may have an optimized Euclidean distance, and mayhave an Euclidean distance that is capable of having gray mapping orcompensating for the Hamming distance, such that MIMO encoding may becarried out using the MIMO matrix at which the parameter ‘a’ isestablished. As described above, the broadcast transmitter may adjustpower of input signals using the MIMO matrix, where the parameter ‘a’may be set to different values according to modulation types of inputsignals.

MIMO encoding may be carried out using the above-mentioned precodingMIMO matrices shown in Equations 12 to 14, and MIMO encoding may becarried out using the phase-rotated MIMO matrices shown in Equations 15to 18. This example may be changed according to designer intention.

A broadcast receiver performs MIMO decoding of the first receptionsignal and the second reception signal, and may generate the firstoutput signal and the second output signal in step S48100. The broadcastreceiver according to one embodiment of the present invention mayperform MIMO decoding using a reverse process of the MIMO encodingprocessed by the broadcast transmitter. Specifically, the broadcastreceiver may synchronize and obtain the index K of the phase rotationused in the above-mentioned matrix, and may calculate LLR using theprecoding matrix.

Thereafter, the broadcast receiver receives the first output signal andthe second output signal, and merges of the first and second outputsignals, such that it can generate the output signal in step S48200.

Modes for Invention

Details about modes for the present invention have been described in theabove best mode.

INDUSTRIAL APPLICABILITY

As described above, the present invention can be wholly or partiallyapplied to digital broadcast systems.

1-4. (canceled)
 5. A method for transmitting broadcast signals in atransmitter, the method comprising: encoding data for Physical LayerPipes (PLPs); mapping the encoded data onto constellations according tomodulation types; Multi-Input Multi-Output (MIMO) processing the mappeddata, wherein a pair of two constellation symbols of the mapped data arecombined based on a rotation matrix with a rotation angle θ, wherein therotation angle θ depends on one of the modulation types, and wherein thecombined symbols are multiplied by a phase rotation matrix; timeinterleaving the MIMO processed data; building frames by mapping thetime interleaved data; modulating the frequency-interleaved data by anOrthogonal Frequency Division Multiplexing (OFDM) scheme; andtransmitting the broadcast signals including the modulated data throughtwo or more antennas.
 6. The method of claim 5, wherein the rotationmatrix is represented as: $\begin{bmatrix}{\cos \; \theta} & {\sin \; \theta} \\{\sin \; \theta} & {{- \cos}\; \theta}\end{bmatrix}\quad$
 7. A method for receiving broadcast signals in areceiver, the method comprising: receiving the broadcast signals throughtwo or more antennas; demodulating the received broadcast signals by anOrthogonal Frequency Division Multiplexing (OFDM) scheme; parsing framesfrom the frequency de-interleaved broadcast signals; timede-interleaving symbols in the parsed frames; Multi-Input Multi-Output(MIMO) decoding the time de-interleaved symbols based on a rotationmatrix with a rotation angle θ and a phase rotation matrix and outputMIMO decoded symbols, wherein the rotation angle θ depends on one ofmodulation types of the MIMO decoded symbols; demapping the MIMO decodedsymbols into data for Physical Layer Pipe (PLP) according to the one ofmodulation types; and decoding the data for the PLP.
 8. The method ofclaim 7, wherein the rotation matrix is represented as: $\begin{bmatrix}{\cos \; \theta} & {\sin \; \theta} \\{\sin \; \theta} & {{- \cos}\; \theta}\end{bmatrix}\quad$
 9. An apparatus for transmitting broadcast signals,the apparatus comprising: an encoder configured to encode data forPhysical Layer Pipes (PLPs); a mapper configured to map the encoded dataonto constellations according to modulation types; a Multi-InputMulti-Output (MIMO) processor configured to MIMO-process the mappeddata, wherein a pair of two constellation symbols of the mapped data arecombined based on a rotation matrix with a rotation angle θ, wherein therotation angle θ depends on one of the modulation types, and wherein thecombined symbols are multiplied by a phase rotation matrix; a timeinterleaver configured to time interleave the MIMO processed data; abuilder configured to build frames by mapping the time interleaved data;a modulator configured to modulate the frequency-interleaved data by anOrthogonal Frequency Division Multiplexing (OFDM) scheme; and atransmitter configured to transmit the broadcast signals including themodulated data through two or more antennas.
 10. The apparatus of claim9, wherein the rotation matrix is represented as: $\begin{bmatrix}{\cos \; \theta} & {\sin \; \theta} \\{\sin \; \theta} & {{- \cos}\; \theta}\end{bmatrix}\quad$
 11. An apparatus for receiving broadcast signals ina receiver, the apparatus comprising: a receiver configured to receivethe broadcast signals through two or more antennas; a demodulatordemodulate the received broadcast signals by an Orthogonal FrequencyDivision Multiplexing (OFDM) scheme; a parser configured to parse framesfrom the frequency de-interleaved broadcast signals; a timede-interleaver configured to time de-interleave symbols in the parsedframes; a Multi-Input Multi-Output (MIMO) decoder configured toMIMO-decode the time de-interleaved symbols based on a rotation matrixwith a rotation angle θ and a phase rotation matrix and output MIMOdecoded symbols, wherein the rotation angle θ depends on one ofmodulation types of the MIMO decoded symbols; a demapper configured todemap the MIMO decoded symbols into data for Physical Layer Pipe (PLP)according to the one of modulation types; and a decoder configured todecode the data for the PLP.
 12. The apparatus of claim 11, wherein therotation matrix is represented as: $\begin{bmatrix}{\cos \; \theta} & {\sin \; \theta} \\{\sin \; \theta} & {{- \cos}\; \theta}\end{bmatrix}\quad$